Low voltage rail to rail high speed analog buffer and method thereof

ABSTRACT

Methods, circuits, and apparatuses that provide Buffer Amplifier, containing Amplifiers and Buffer Drivers, one or more of the following: ultra low power Buffer Amplifier, capable of having high gain, low noise, high speed, near rail-to-rail input-output voltage span, high sink-source current drive capability for an external load, and able to operate at low power supply voltages. Methods, circuits, and apparatuses that provide regulated cascode (RGC) current mirrors (CM) capable of operating at low power supply and having wide input-output voltage spans.

CROSS-REFERENCE TO RELATED PATENT APPLICATIONS

This application claims the priority benefit of U.S. patent applicationSer. No. 15/451,334, filed Mar. 6, 2017 and entitled Ultra Low PowerHigh-Performance Amplifier”; U.S. Provisional Patent Application Ser.No. 62/304,373, filed Mar. 7, 2016 and entitled “Class AB amplifier lowpower and fast”; U.S. Provisional Patent Application Ser. No. 62/320,512filed Apr. 9, 2016 and entitled “Class AB amplifier high gain”; and U.S.Provisional Patent Application Ser. No. 62/415,496 filed Oct. 13, 2016and entitled “Class AB Amplifier Low Noise”; Each of the aboveapplications are herein specifically incorporated by reference in theirentirety.

FIELD OF DISCLOSURE

This disclosure related to improvements in current mirrors, currentsources, amplifiers, and output buffer drives for use in integratedcircuits (ICs).

BACKGROUND

Operating ICs under ultra low currents and low power supplies, incomplementary metal-oxide semiconductor (CMOS) technology, pose seriouschallenges in the design of integrated circuits. Low operating currentscause lower speeds and lower gain and higher noise in an IC. Also,rail-to-rail operations for ICs becomes a necessity given thatsignal-to-noise requirements at low power supplies demand input andoutput terminals of ICs to get as close as possible to the powersupplies.

SUMMARY OF THE DISCLOSURE

An aspect of the present disclosure is to make small and low costcurrent sources, current mirrors, amplifiers, buffers drivers, andbuffer amplifiers that can operate with one or more of the followingcharacteristics: (a) wide input-output voltage span, (b) low powersupply voltage, (c) low power consumption, (d) low noise, (e) fastdynamic response, (f) symmetric design to minimize systematic errors,(g) use simple design that generally improves performance tospecifications over operating and process variation, and/or (h) make theIC rugged for long term manufacturing using standard CMOS fabricationprocess that is inexpensive, and readily available at multiplefabrication factories, thus easing process node portability.

Another aspect of the disclosure includes a current mirror with highimpedance, wide input-output span, and low drain voltage (V_(DD)VDD).Another aspect of this disclosure is to make current sources and currentmirrors with high output impedance, wide input-output voltage range, andoperating with low power supply voltage. This is accomplished by amethod of making a ‘current source’ or a ‘current mirror’ comprisingregulated cascode current mirror (RGC-CM) coupled with diode connectedself cascode (DCSC). This may also be accomplished by another method ofmaking a ‘current source’ or a ‘current mirror’ comprising RGC-CM couplewith inverting current mirror amplifier (ICMA). Moreover, this goal ismet by another method of making a ‘current source’ or a ‘current mirror’comprising RGC-CM coupled with a composite amplifier (CSGA) thatcontains common source amplifier (CSA) and common gate amplifier (CGA)with the gate of CGA connected to source of CSA.

Another aspect of the disclosure further includes an amplifier using thedisclosed current mirrors discussed above to make amplifiers with highgain and wide input-output range, and operating with low power supplyvoltage. This is accomplished by a method of making an ‘amplifier’comprising plurality of the disclosed RGC-CM couple with DCSC. This mayalso be accomplished by another method of making an ‘amplifier’comprising plurality of the disclosed RGC-CM coupled with ICMA.Moreover, this goal is also met by method of making an ‘amplifier’utilizing plurality of the disclosed RGC-CM coupled with a compositeamplifier (CSGA).

Another aspect of the disclosure further includes a floating currentsource with low V_(DD)VDD that also operates fast. Another aspect ofthis disclosure is to emulate the function of a floating current sources(FCS) that equalizes upper and lower current sources and is capable ofoperating with low power supply voltages. This may be accomplished by amethod utilizing cascoded PMOSFETs (e.g., one PMOSFET on top and secondPMOSFET in the middle), and cascoded NMOSFETs (e.g., one NMOSFET on topand second NMOSFET in the middle). The middle PMOSFET and the middleNMOSFET drain and source currents are crisscrossed and fed to oneanother. A lower regulating circuit holds the VGS of the middle NMOSFETconstant by regulating the gate voltage of the lower NMOSFET. An upperregulating circuit holds the VGS of the middle PMOSFET constant byregulating the gate voltage of the upper PMOSFET. As a result the netsum of current in the upper PMOSFET and lower NMOSFET is equalized.

A further aspect of this disclosure is an amplifier using the disclosedFCS so that the amplifier's upper and lower current sources areequalized in order to improve amplifier's performance. Some of suchimprovements are in reducing the amplifier's offsets due to currentasymmetries when the amplifier's input has a wide common mode span,while enabling the amplifier to operate with low power supply voltage.

A further aspect of this disclosure is reducing an amplifier's outputnoise, when the amplifier consumes low currents, operates at lowV_(DD)VDD, while the amplifier has wide input-output voltage span, andit is fast. Another aspect of this disclosure is to reduce anamplifier's noise by narrow banding it. When the amplifier's inputs areimbalanced in response to a large transient input signal, to make up forlost speed due to narrow-banding the amplifiers, the operating currentof the amplifiers is dynamically and rapidly boosted to I_(Peak)=h×I_(Q)(in the current boost-on phase) where I_(Q) is the steady stateoperating current of the amplifier set at low current levels to savepower consumption. After the amplifier's inputs approach steady stateand are near balance (not precisely equal, just roughly in balance), theamplifier's operating current is dynamically and slowly decreased(current boost-off phase) back to I_(Q) levels. The amplifier dynamicresponse is improved by enhancing both the Slew Rate (SR) and thesettling time (τ_(s)) of the amplifier. In summary, to rejovinate the(narrow banded) amplifier's SR while its' inputs are imbalanced due toreceiving a large-signal transient, the amplifiers' operating currentreceive a very fast I_(Peak) pulse during the current boost-on phase.Moreover, to rejovinate the (narrow banded) amplifier's τ_(s) when theamplifier's inputs approach balance and amplifiers enters the boost-offphase, instead of shutting off I_(Peak) very fast, the amplifier'soperating current starts decaying slowly (following a time constant thattracks the amplifier's AC response) from I_(Peak) towards I_(Q).Accordingly, when the amplifier operating current is at I_(Q)equilibrium or steady state, the amplifier's noise is reduced since itis narrow banded.

Another aspect of the disclosure herein is a buffer driver that operateswith low V_(DD)VDD, has wide input output voltage span, and is fast byoperating chiefly in current mode. A further aspect of this disclosureis an amplifier using the disclosed buffer driver that can operate withlow V_(DD), have near rail-to-rail input-outputs spans, and mostlyoperate in current mode which improve speed at low currents. This isaccomplished by a method of making an ‘amplifier’ comprising pluralityof minimum current selectors (MCS) or loser take all (LTA) which,directly or indirectly, monitor the sink-source currents of the bufferdriver's output field effect transistors (FETs). Concurrently, anon-inverting current mirror amplifier (NICMA) or inverting currentmirror amplifier (ICM) or inverting current feedback amplifiers (ICFA)would regulate the minimum stand-by currents for either the inactivesink output transistor or the inactive source output transistors. Also,in order to lower the current consumption associated with monitoring thesink source output transistor currents, a complementary non-invertingcurrent mirror (CNICM) is utilized to (rectify) curb the sink-sourcesignals before they are fed to the MCS or LTA.

Aspects of the embodiments disclosed herein further include that theycan often be fabricated in standard digital CMOS; and embodiments havesmall size (e.g., for low cost and high volume applications); and theembodiments typically operate MOSFETs in subthreshold so they canoperate at ultra low currents and low power supply voltages needed inparticular to emerging wireless and battery less applications.

Aspects of the embodiments disclosed herein further include a method ofoperating an ultra low power Buffer Amplifier (BA), containing anAmplifier coupled with a Buffer Driver (BD), comprising: increasing gainand widening the input-output voltage span of the Amplifier utilizing aplurality of regulated cascode (RGC) current mirrors (RGC-CM) where eachRGC-CM is made of at least one of the following three circuits: 1) adiode connected self cascode (DCSC) coupled with a common sourceamplifier (CSA); 2) a current mirror amplifier (CMA), inverting ornon-inverting type, that contains a common source amplifier (CSA); and3) a CSA coupled with one common gate amplifier (CGA) wherein the commongate terminal of the CGA is connected to the common source terminal ofthe CSA; lowering the minimum operating power supply and reducing offsetof the Amplifier by utilizing a current equalizer circuit that emulatesthe function of a floating current source (FCS) having at least twocomplementary cascoded current sources made of field effect transistors(FETs), wherein middle cascoded FET's gate to source voltages (VGS) areheld constant by regulating the VGS of the lower FETs, whose currentsare equalized and mirrored into the Amplifier's bias network, where thelower FET's source terminals are connected to the power supplies;lowering output noise of the Amplifier, by narrow banding theAmplifier's high gain node, while concurrently rejuvenating the narrowbanded Amplifier's speed by utilizing one of the following circuits: 1)minimum current select (MCS); and 2) loser take all (LTA) circuits togenerate dynamically boosted operating current when the Amplifier'sinputs are intermittently imbalanced; and lowering the minimum operatingpower supply, having near rail-to-rail input output voltage span, havinghigh-speed, providing large sink-source current for output load, whileregulating the operating current in the inactive sink-source transistorof the Buffer Amplifier (BA) by utilizing a Buffer Driver (BD) thatcontains at least one of the following circuits: 1) MCS; 2) LTA; 3)current mirror amplification (CMA); and 4) complementary current mirrorthat enable the Buffer Driver (BD) to chiefly operate and processsignals in current mode.

Aspects of the embodiments disclosed herein further include a method ofreducing output noise in an amplifier comprising: narrow banding theamplifier to keep the amplifier's static current consumption low;rejuvenating the dynamic response of the narrow banded amplifier bydynamically boosting the amplifier's operating current when theamplifier's inputs receive a large transient signal that cause animbalance at the amplifier's inputs; and returning back to the steadystate conditions when the amplifiers inputs are substantially equalizedand shutting off the dynamic boosting of the amplifier's operatingcurrent when the amplifier's current consumption returns to low levelsand the amplifier's output noise is reduced. The method furthercomprising: connecting a first capacitor to the high impedance or highgain node of the amplifier to narrow band the amplifier; and wherein thefirst capacitor can be an active or a passive capacitor that isintrinsic or extrinsic at the high impedance or high gain node of theamplifier. The method further comprising: rejuvenating the dynamicresponse of the narrow banded amplifier by speeding up the amplifier'sslew rate and settling time upon detecting an imbalance at theamplifier's inputs by utilizing one of a: 1) a loser take all (LTA)circuit; and 2) minimum current selector (MCS) circuit; applying theoutput of the LTA circuit or MCS circuit to generate a ‘boost on’ or‘boost off’ signal; using the ‘boost on’ signal to boost the operatingcurrent of the amplifier; and using the ‘boost off’ signal to return theamplifier's operating current back to the low static current at steadystate condition and attaining lower output noise for the amplifier atsteady state conditions. The method further comprising: using the ‘booston’ signal to rapidly boost the operating current of the amplifier toincrease the amplifier's slew rate (SR); using the ‘boost off’ signal togenerate a slow declining current, with a slow decay to zero to speed upthe settling time (τ_(s)) of the amplifier; improving the dynamicresponse of the amplifier by optimizing both its SR and τ_(s); and usingthe ‘boost off’ signal to return the amplifier's operating current backto the low static current at steady state condition and attaining lowoutput noise for the amplifier during steady state conditions.

Aspects of the embodiments disclosed herein further include a bufferdriver circuit comprising: first output driver field effect transistors(FETs) having the function of sinking and sourcing currents for anexternal load; at least one of a minimum current selector (MCS) signalor at least one loser take call (LTA) signal having the function ofmonitoring and processing the sink-source currents of the first outputdriver FETs; at least one of a group consisting of a: 1) current mirroramplifier (CMA); 2) an inverting CMA (ICMA); and 3) non-inverting CMA(NICMA) having the function of receiving the MCS or LTA signals, andutilizing the MCS or LTA signals for regulating and controlling thecurrent in the inactive sink-source FET; a first buffer driver thatutilizes at least one MCS or LTA signals and at least one of the CMA,ICMA and NICMA; and wherein the buffer driver has the function ofsinking and sourcing current for external loads and regulating a minimumoperating current in the inactive sinking or sourcing FET. The circuitfurther comprising: one of the group consisting of: 1) a firstcomplementary non-inverting current mirror (CNICM); and 2) a firstcomplementary inverting current mirror (CICM) having a function ofmonitoring the sink-source output driver FETs current and generatingrectified sink-source signals; and providing the rectified sink-sourcesignals to the minimum current selector (MCS) or loser take call (LTA)circuits to process the sink-source output driver FETs signals. Thecircuit capable of being used in a first amplifier further comprising:wherein the first amplifier utilizes the first buffer driver circuit inorder for the first amplifier to be coupled with the buffer driver to becapable of sinking and sourcing current for external loads andregulating a minimum operating current in the inactive sinking orsourcing FET.

Aspects of the embodiments disclosed herein further include a method ofoperating at least one regulated cascode (RGC) current mirror (RGC-CM)comprising: supplying voltage to the at least one RGC-CM from a positivesupply voltage (V_(DD)) and a negative supply voltage (V_(SS));increasing the output resistance of the at least one RGC-CM by a firstauxiliary amplifier; and widening the voltage span of the input-outputterminals of the at least one RGC-CM by generating a direct current (DC)voltage shift from at least one diode connected self cascode (DCSC)coupled to the first auxiliary amplifier. The method further comprising:utilizing a first amplifier that contains a plurality of RGC-CMs,wherein each of the plurality of RGC-CMs are utilized in the firstamplifier to function as current mirrors; delivering power to the firstamplifier by a positive supply voltage (V_(DD)) and a negative supplyvoltage (V_(SS)); increasing the gain of the first amplifier byutilizing the plurality of RGC-CMs; and widening the input-output spanof the first amplifier by utilizing the plurality of RGC-CMs.

Aspects of the embodiments disclosed herein further include at least oneregulated cascode (RGC) current mirror (RGC-CM) circuit comprising:cascoded transistors having a first transistor placed in series at afirst node with a second transistor wherein the output of the RGC-CM isthe drain terminal of the second transistor; a first diode connectedself cascode (DCSC) having a diode connected third transistor in serieswith a fourth transistor, wherein the gate terminals of the thirdtransistor and the fourth transistor are connected together and whereinthe source of the third transistor is connected to the drain of thefourth transistor at the second node, wherein the source of the fourthtransistor is connected to the first node; and a first auxiliaryamplifier (AA) whose input is connected to the second node, and whereinthe output of the first AA is connected to the gate terminal of thesecond transistor. The circuit utilized in a first amplifier furthercomprising: a plurality of the at least one regulated cascode (RGC)current mirrors (RGC-CMs); wherein each of the plurality of RGC-CMsfunction as current mirrors; wherein the gain of the first amplifier isincreased by utilizing the plurality RGC-CMs; and wherein theinput-output span of the first amplifier is widened by utilizing theplurality of RGC-CMs.

Aspects of the embodiments disclosed herein further include a method tooperate a regulated cascode (RGC) current mirror (RGC-CM) comprising:providing power to the RGC-CM by a positive supply voltage (V_(DD)) anda negative supply voltage (V_(SS)); operating input and output signalsin a current mode by utilizing a first inverting current mode amplifier(ICMA) that includes a first inverting current mirror (ICM); generatingamplification through the first ICMA to perform the function of a firstauxiliary amplifier (AA); increasing the output resistance of the RGC-CMby utilizing the first ICMA; and widening the input-output terminalspans of the RGC-CM by utilizing the first ICMA. The method ofcomprising: utilizing a first amplifier having a plurality of regulatedcascode (RGC) current mirrors (RGC-CMs) wherein each of the plurality ofRGC-CMs are utilized in the first amplifier function as current mirrors.

A regulated cascode (RGC) current mirror (RGC-CM) circuit comprising:cascoded transistors having a first transistor placed in series at afirst node with a second transistor wherein the output of the RGC-CM isthe drain terminal of the second transistor; a first current mirror (CM)with a diode connected third transistor and a fourth transistor; a firstcurrent mirror amplifier (CMA) having the first current mirror, and athird transistor whose source terminal, which is the input of CMA, isconnected to the first node and its drain terminal is connected to thegate and drain terminals of the third transistor; wherein the output ofCMA, which is the drain terminal of the fourth transistor, is connectedto the gate terminal of second transistor; and wherein the CMA functionsas the auxiliary amplifier for the RGC-CM in order to increase theoutput resistance of the RGC-CM. The circuit used in a first amplifierfurther comprising: wherein each of the plurality of RGC-CMs that areutilized in the first amplifier function as current mirrors; wherein thegain of the first amplifier is increased by utilizing the pluralityRGC-CMs; and wherein the input-output span of the first amplifier iswidened by utilizing the plurality of RGC-CM.

Aspects of the embodiments disclosed herein further include a regulatedcascode (RGC) current mirror (RGC-CM) circuit comprising: cascodedtransistors having a first transistor placed in series at a first nodewith a second transistor wherein the output of the RGC-CM is the drainterminal of the second transistor; a first common source amplifier(CSA); a first common gate amplifier (GGA); wherein the inputs of thefirst CSA are connected to the source terminal of the first transistorand the first node; wherein the output of the first CSA is connected tothe input of the first CGA; wherein the common gate terminal of thefirst CGA is connected to the common source terminal of the first CSA;and wherein the first CGA's output terminal is connected to the gate ofthe second transistor. The circuit used in a first amplifier furthercomprising: a plurality of regulated cascode (RGC) current mirrors(RGC-CMs); and wherein each of the plurality of RGC-CMs that areutilized in the first amplifier function as current mirrors; wherein thegain of the first amplifier is increased by utilizing the pluralityRGC-CMs; and wherein the input-output span of the first amplifier iswidened by utilizing the plurality of RGC-CM.

Aspects of the embodiments disclosed herein further include a regulatedcascode (RGC) current mirror (RGC-CM) circuit comprising: cascodedtransistors having a first transistor placed in series at a first nodewith a second transistor wherein the output of the RGC-CM is the drainterminal of the second transistor; a first common source amplifier(CSA), with a built in offset, having a third transistor and a fourthtransistor, whose source terminals are connected to the second node,wherein the gate of the third transistor which is one of the inputs ofCSA is connected to the first node, wherein the other terminal of theCSA, which is the gate and drain terminals of the fourth transistor areconnected together and are connected to the source terminal of the firsttransistor; and a first common gate amplifier (CGA), comprising a fifthtransistor whose gate terminal is connected to the second node, wherethe input of the CGA which is the source terminal of is connected to thefirst node, and the output of the CGA which is the drain terminal of thefifth transistor is connected to the gate terminal of the secondtransistor. The circuit used in a first amplifier, further comprising: aplurality of regulated cascode (RGC) current mirrors (RGC-CMs), whereineach of the plurality of RGC-CMs that are utilized in the firstamplifier function as current mirrors; wherein the gain of the firstamplifier is increased by utilizing the plurality RGC-CMs; and whereinthe input-output span of the first amplifier is widened by utilizing theplurality of RGC-CMs.

Aspects of the embodiments disclosed herein further include a currentequalizing circuit comprising: a positive supply voltage (V_(DD)) and anegative supply voltage (V_(SS)); a first Positive Metal Oxide SiliconField Effect Transistor (PMOSFET) and second PMOSFET forming a cascodedcurrent source, wherein the drain of the first PMOSFET is connected tothe source of the second PMOSFET at the first node; a first NegativeMetal Oxide Silicon Field Effect Transistor (NMOSFET) and second NMOSFETforming a cascoded current source, wherein the drain of the firstNMOSFET is connected to the source of the second NMOSFET at the secondnode; wherein the second PMOSFET drain terminal is connected to thesource terminal of and second NMOSFET at the second node; wherein thesecond NMOSFET drain terminal is connected to the source terminal of andsecond PMOSFET at the first node; wherein a first regulating circuitkeeps the gate-to-source voltage of the second PMOSFET substantiallyconstant by regulating the gate-to-source voltage of the first PMOSFET;wherein a second regulating circuit keeps the gate-to-source voltage ofthe second NMOSFET substantially constant by regulating thegate-to-source voltage of the first NMOSFET; and wherein the current inthe first PMOSFET and first NMOSFET are substantially equalized. Thecircuit used in a first amplifier further comprising: the firstamplifier including the first current equalizing circuit of claim 22.

Aspects of the embodiments disclosed herein further include a currentequalizing method comprising: delivering power to the current equalizerby a positive supply voltage (V_(DD)) and a negative supply voltage(V_(SS)); lowering the minimum operating power supply and reducingoffset of the Amplifier (A) by utilizing a current equalizer circuitthat emulates the function of a floating current source (FCS) containingtwo complementary cascode current sources, wherein the cascoded fieldeffect transistors (FETs) gate to source voltages (VGS) are heldconstant by regulating the VGS of the lower FETs, whose currents areequalized and mirrored into the amplifier's bias network, and whosesource terminals are connected to the power supplies. Aspects of theembodiments disclosed herein further include a method of regulatingminimum operating current in a sink transistor and a source transistorof a buffer driver in an integrated circuit comprising: monitoring asink transistor current of a sink transistor and a source transistorcurrent of a source transistor in a buffer driver; detecting animbalance between the sink transistor current and the source transistorcurrent; generating a regulation signal in response to detecting theimbalance between the sink transistor current and the source transistorcurrent by utilizing a current feedback amplifier (CFA); and utilizingthe regulation signal to regulate a minimum operating current in atleast one of the sink transistor and the source transistor. Aspects ofthe embodiments disclosed herein further include a buffer driver circuitcomprising: a buffer driver having a buffer driver input port and abuffer driver output port; a driver field effect transistor (FET) havinga driver FET input port and a driver FET output port, a currentimbalance detection (CID) circuit having a CID circuit input port and aCID circuit output port; a current amplifier (CA) circuit having a CAcircuit input port and a CA circuit output port; wherein the driver FEToutput port is capable of communicating with the buffer driver outputport; wherein the driver FET input port is capable of communicating withthe buffer driver input port; wherein the CID circuit input port iscapable of communicating with the driver FET input port and the bufferdriver input port; wherein the CID circuit output port is capable ofcommunicating with the CA circuit input port; and wherein the CA circuitoutput port is capable of communicating with the driver FET, the CIDcircuit input port, and the buffer driver input port.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1A is a schematic circuit diagram of the embodiment illustrating anamplifier utilizing FIG. 1B.

FIG. 1B is a schematic circuit diagram of the embodiment illustrating acurrent mirror utilizing a regulated cascode current mirror (RGC-CM)coupled with diode connected self cascode (DCSC)

FIG. 1C is a schematic circuit diagram of the embodiment illustrating anamplifier utilizing FIG. 1D.

FIG. 1D is a schematic circuit diagram of the embodiment illustrating acurrent mirror utilizing a RGC-CM coupled with an inverting currentmirror amplifier (ICMA).

FIG. 1E is a schematic circuit diagram of the embodiment illustrating anamplifier utilizing FIG. 1F.

FIG. 1F is a schematic circuit diagram of the embodiment illustrating acurrent mirror utilizing a RGC-CM coupled with a composite amplifier(CSGA).

FIG. 1G is a schematic circuit diagram of an embodiment illustrating aprior art RGC-CM.

FIG. 1H is a circuit simulation showing a frequency response of theamplifier illustrated FIG. 1A.

FIG. 1I is a circuit simulation showing a transient response and currentconsumption of the amplifier illustrated FIG. 1A.

FIG. 1J is a circuit simulation showing a frequency response of theamplifier illustrated FIG. 1C.

FIG. 1K is a circuit simulation showing a transient response and currentconsumption of the amplifier illustrated FIG. 1C.

FIG. 1L is a circuit simulation showing a frequency response of theamplifier illustrated FIG. 1E.

FIG. 1M is a circuit simulation showing a transient response and currentconsumption of the amplifier illustrated FIG. 1E.

FIG. 2A is a schematic circuit diagram of the embodiment illustrating anamplifier utilizing the first floating current source (FCS200A)embodiment

FIG. 2B is a schematic circuit diagram of the embodiment illustrating anamplifier utilizing the second floating current source (FCS200B)embodiment

FIG. 2C is a schematic circuit diagram of the embodiment illustrating anamplifier utilizing the third floating current source (FCS200C)embodiment

FIG. 2D is a schematic circuit diagram of a prior art floating currentsource (FCS) embodiment

FIG. 2E is a circuit simulation showing transient response, currentconsumption, and change in the FCS200A current in response to change ininput-output voltage of the amplifier illustrated FIG. 2A.

FIG. 2F is a circuit simulation showing transient response, currentconsumption, and change in the FCS200B current in response to change ininput-output voltage of the amplifier illustrated FIG. 2B.

FIG. 2G is a circuit simulation showing transient response, currentconsumption, and change in the FCS current in response to change ininput-output voltage of the amplifier illustrated FIG. 2C.

FIG. 3A is a schematic circuit diagram of the embodiment illustrating anamplifier utilizing a first noise reduction and speed boost circuit.

FIG. 3B is a schematic circuit diagram of the embodiment illustrating anamplifier utilizing a second noise reduction and speed boost circuit.

FIG. 3C is a circuit simulation showing output noise of the amplifierillustrated in FIG. 3A with and without the noise reduction & speedboost.

FIG. 3D is a circuit simulation showing transient response, currentconsumption of the amplifier illustrated in FIG. 3A, with and withoutthe noise reduction & speed boost.

FIG. 4A is a schematic circuit diagram of a prior amplifier's gain stagewith near rail-to-rail input and output voltage span, without a bufferdriver.

FIG. 4B is a schematic circuit diagram of the embodiment illustrating abuffer driver utilizing complementary non-inverting current mirror(CNICM), minimum current selector (MCS), and inverting current mirroramplifier (ICMA).

FIG. 4C is a schematic circuit diagram of the embodiment illustrating abuffer driver utilizing complementary non-inverting current mirror(CNICM), loser take all (LTA), and non-inverting current mirroramplifier (NICMA).

FIG. 4D is a schematic circuit diagram of the embodiment illustrating abuffer driver utilizing minimum current selector (MCS), and invertingcurrent mirror amplifier (ICMA).

FIG. 4E is a circuit simulation showing the sink-source output FETcurrents and I_(DD) as a function of external resistive load for anamplifier containing the gain stage of FIG. 4A coupled with bufferdriver illustrated in FIG. 4B.

FIG. 4F is a circuit simulation showing sink-source output FET currentsand I_(DD) as a function of external resistive load for an amplifiercontaining the gain stage of FIG. 4A coupled with buffer driverillustrated in FIG. 4C.

FIG. 4G is a circuit simulation showing sink-source output FET currentsand I_(DD) as a function of external resistive load for amplifiercontaining the gain stage of FIG. 4A coupled with buffer driverillustrated in FIG. 4D.

FIG. 5 is a block level diagram showing the Buffer Amplifier having theAmplifier coupled with the Buffer Driver section.

DETAILED DESCRIPTION Definitions, Acronyms and Abbreviations

The following terms, definitions, acronyms, term usages andabbreviations are explained below and used throughout this application:

TERMS DESCRIPTION IC or chips Integrated circuits Amplifier Has highinput and high output impedance and the amplifier amplifies signalsapplied to it its input, unless otherwise specified buffer driver Hashigh input impedance and low output impedance with high current drivecapability so the buffer driver can sink and source currents for anexternal load, unless otherwise specified buffer amplifier Includes anamplifier coupled with a buffer driver, combination of which can amplifysignals applied to the buffer amplifier inputs. Buffer amplifier hashigh input impedance, and low output impedance with high output currentdrive capability so it can sink and source currents for an externalload, unless otherwise specified V_(DD) or positive rail Positive powersupply voltage V_(SS) or negative rail Negative power supply voltageRail-to-rail input and output Input and output terminals spanning(substantially near) to V_(DD) and V_(SS) GND ground voltage, which canbe the same as V_(SS) V_(IN) Input voltage containing the negativeterminal V_(IN−) and the positive terminal V_(IN+) VIN_(CM) Common modeinput voltage V_(OUT) output voltage Bias current or reference currentUsed interchangeably or current source I_(DD) Current flowing throughV_(DD) or current consumption FET Field Effect Transistors JFET JunctionField Effect Transistors BJT Bipolar Junction Transistors CMOSComplementary Metal Oxide Semiconductor (may be operated in subthresholdor normal regions) BiCMOS BJT and CMOS transistors fabricated on thesame water MOSFET Metal Oxide Semiconductor Field Effect Transistors(may be operated in subthreshold or normal regions) MOSFET's linear ortriode or Used interchangeably resistive regions of operationsSubthreshold MOSFET's Subthreshold region of operation CMOS, PMOS, andNMOS or Used interchangeably CMOSFET, PMOSFET, and NMOSFET, respectivelyGate, source, and drain or gate Used interchangeably terminals, sourceterminal, and drain terminals respectively W/L of a MOSFET Width overlength ratio (aspect ratio) of MOSFETs W/L of a MOSFET or aspect Usedinterchangeably ratios Ohms per square or Ω/square Resistivity ofmaterial per square area Terms applied to the W/L of Usedinterchangeably MOSFETs such as ‘predetermined’ versus ‘programmed’versus ‘set’ C. or ° C. Used interchangeably and shows unit oftemperature in Celsius Ω Ohms, unit of measurement for resistivity AAmpere V Volt Mm Micro-meter or 10⁻⁶ meter (e.g., W = 4 μm or W = 4 ×10⁻⁶ meter N Nano or 10⁻⁹ (e.g., nA = nano ampere or 10⁻⁹ A) P Pico or10⁻¹² (e.g., pF = pico Farad or 10⁻¹² F M MOSFET carrier mobility V_(T)Thermal voltage C_(OX) Gate oxide capacitance of a MOSFET C_(e)Effective capacitance Ce_(MXXXX) Effective input gate terminalcapacitance of a MOSFET (e.g., Ce_(M316A) is the effective capacitancelooking into the gate of FET M_(316A)) Ce_(xxxx) Effective capacitanceat node xxxx (e.g., Ce_(314A) is the effective capacitance at node 314A)V_(TH) Threshold voltage of a MOSFET V_(A) or 1/λ Used interchangeablyto show MOSFET early voltage β Beta = β = μ × C_(OX) to show a MOSFETgain M To show MOSFET mobility exponent factor used in device equationsH To show MOSFET subthreshold slope factor used in device equationsV_(OFS) Offset voltage between two MOSFETs, mostly referring to a CMOSamplifier's input offset voltage V_(xxxx) Voltage at node xxxx (e.g.,V_(108A) is the voltage at node 108A) V_(DS) or VDS_(sat) or V_(on) Usedinterchangeably is the drain terminal voltage to source terminal voltageof a MOSFET, including in the saturation region V_(GS) Gate terminalvoltage to source terminal voltage of a MOSFET VG_(Mxxx) Voltage at thegate terminal of a MOSFET (e.g., VG_(M128D) is the voltage at the gateterminal of M_(128D)) VD_(Mxxxx) Voltage at the drain terminal of aMOSFET (e.g., VD_(M128D) is the voltage at the drain terminal ofM_(128D)) VS_(Mxxxx) Voltage at the source terminal of a MOSFET (e.g.,VS_(M128D) is the voltage at the source terminal of M_(128D))VGS_(Mxxxx) Gate to source voltage of a MOSFET M_(xxxx) (e.g.,VGS_(M128D) is V_(GS) of M_(128D)) VDS_(Mxxxx) Drain to source voltageof a MOSFET M_(xxxx) (e.g., VDS_(M128D) is V_(DS) of M_(128D)) ΔV_(GS)Difference between the gate to source voltage of two MOSFETs Δv_(O), orΔv_(OUT) Change in an output voltage of a function (e.g., amplifier'soutput or current mirror's output) Δv_(X) Small change in a voltageΔi_(X) Small change in a current I_(DS) or I_(D) or ID_(Mxxxx) orI_(Mxxx) Used interchangeably and it is the drain to source current of aMOSFET (e.g., I_(M117D) is I_(DS) of M_(117D)) I_(xxxx) It is a currentsource (e.g., I_(107B) is the current source #107B) g_(m) or gm_(MOSFET)To show transconductance of MOSFET where g_(m) ≈ I_(D)/V_(T) for MOSFEToperating in subthreshold (ignoring second order effects, such as η orV_(TH) or V_(A) for clarity of description) r_(O) or r_(ds) Outputresistance of MOSFET or ∝ V_(A)/I_(D) for MOSFET operating insubthreshold R_(OUT) or R_(O) Used interchangeably and shows the outputresistance of a function such as that of a current source, currentmirror, transconductance amplifier, or an auxiliary amplifier Ze_(xxxx)Effective impedance or effective resistance at node xxxx (e.g.,Ze_(314A) is the effective impedance at node 314A) TC Temperaturecoefficient AC response Small signal frequency output versus inputresponse of a circuit A_(V) Main amplifier's open loop gain at DC F_(U)Unity gain frequency of amplifier (stated in Hz = Hertz) P_(M) Phasemargin of amplifier at F_(U) (stated in ° = degrees) SR Slew rate ofamplifier (stated in V/μS or volts per micro seconds) t_(S) Settlingtime of amplifier's V_(OUT) to 5% of steady state (stated in μS = microseconds) PSRR Power supply rejection ratio of amplifier ‘a’ through ‘z’To show the ratio of MOSFET W/Ls or their aspect ratios used in acircuit design ~ or ≈ Approximately equal « Significantly less than(e.g., ID_(M304) « 2i Means ID_(M304) is significantly less than 2i andID_(M304) can be zero or substantially close to it ≈ Approximately equal∝ As a function of or proportional to ⇒ or → Implication, results SCMOSFET self-cascode DCSC or DCSC_(xxxx) MOSFET diode connected selfcascode or DCSC number xxxx RGC or RGC_(xxxx) Regulated Cascode or RGCnumber xxxx CM or CM_(xxxx) current mirror or CM number xxxx RGC-CM orRGC-CM_(xxxx) Regulated Cascode current mirror or RGC-CM number xxxRGC-CS Regulated Cascode current source RGC or RGC-CM or RGC-CS Usedinterchangeably, unless otherwise noted A_(xxxxx) Amplifier used in aregulating circuit (e.g., A_(P200C) is an amplifier number P200C withPMOSFET inputs) A_(AUX) or AA Auxiliary amplifier used for example inregulate cascode current mirror (RGC-CM) MCS or MCS_(xxxx) Minimumcurrent selector or MCS number xxxx Buffer or Buffer Driver or Usedinterchangeably for buffer driver or BUF number BUF_(xxxx) xxxx Min(ID_(Mxxxx), ID_(Myyyy)) Selecting minimum of 2 FET currents (e.g. whenID_(M412D) < ID_(M414D) then Min (ID_(M412D), ID_(M414D)) = ID_(M412D)LTA or LTA_(xxxx) Loser take all or LTA number xxxx LTA (ID_(xxxxx),ID_(yyyyy)) Similar to MCS, the loser of 2 FET currents takes all (e.g.when ID_(M412D) < ID_(M414D) then LTA (ID_(M412D), ID_(M414D)) =ID_(M412D) WTA or WTA_(xxxx) Winner take all LTAA or LTAA_(xxxx) Losertake all amplifier or LTAA number xxxx ICM or ICM_(xxxx) Invertingcurrent mirror or ICM number xxxx NICM or NICM_(xxxx) Non-Invertingcurrent mirror or NICM number xxxx ICMA or ICMA_(xxxx) Inverting currentmirror amplifier or ICMA number xxxx NICMA or NICMA_(xxxx) Non-Invertingcurrent mirror amplifier or NICMA number xxxx FCTA Folded cascodetransconductance amplifier FCA, or folded cascode Used interchangeablyamplifier, or cascode amplifier, or transconductance amplifier FCS orFCS_(xxxx) Floating current source or FCS number xxxx CSA or CSA_(xxxx)Common source amplifier or CSA number xxx CGA or CSA_(xxxx) Common gateamplifier or CSA number xxxx KCL Kirchhoff's Current Law KVL Kirchhoff'sVoltage Law

Numerous embodiments are described in the present application and arepresented for illustrative purposes only and is not intended to beexhaustive. The embodiments were chosen and described to explainprinciples of operation and their practical applications. The presentdisclosure is not a literal description of all embodiments of thedisclosure(s). The described embodiments also are not, and are notintended to be, limiting in any sense. One of ordinary skill in the artwill recognize that the disclosed embodiment(s) may be practiced withvarious modifications and alterations, such as structural, logical, andelectrical modifications. For example, the present disclosure is not alisting of features which must necessarily be present in allembodiments. On the contrary, a variety of components are described toillustrate the wide variety of possible embodiments of the presentdisclosure(s). Although particular features of the disclosed embodimentsmay be described with reference to one or more particular embodimentsand/or drawings, it should be understood that such features are notlimited to usage in the one or more particular embodiments or drawingswith reference to which they are described, unless expressly specifiedotherwise. The scope of the disclosure is to be defined by the claims.

Although process (or method) steps may be described or claimed in aparticular sequential order, such processes may be configured to work indifferent orders. In other words, any sequence or order of steps thatmay be explicitly described or claimed does not necessarily indicate arequirement that the steps be performed in that order. The steps ofprocesses described herein may be performed in any order possible.Further, some steps may be performed simultaneously despite beingdescribed or implied as occurring non-simultaneously (e.g., because onestep is described after the other step). Moreover, the illustration of aprocess by its depiction in a drawing does not imply that theillustrated process is exclusive of other variations and modificationsthereto, does not imply that the illustrated process or any of its stepsare necessary to the embodiment(s). In addition, although a process maybe described as including a plurality of steps, that does not imply thatall or any of the steps are essential or required. Various otherembodiments within the scope of the described disclosure(s) includeother processes that omit some or all of the described steps. Inaddition, although a circuit may be described as including a pluralityof components, aspects, steps, qualities, characteristics and/orfeatures, that does not indicate that any or all of the plurality areessential or required. Various other embodiments may include othercircuit elements or limitations that omit some or all of the describedplurality.

Throughout this disclosure, the body terminal of PMOSFETs can be eitherconnected to their respective PMOSFET source terminals or to thepositive power supply, V_(DD). Similarly here, the body of NMOSFETs canbe either connected to their respective NMOSFET source terminal or tothe negative power supply, V_(SS). Moreover, the negative supplyvoltage, V_(SS), can be alternatively connected to the ground (GND)potential. Given that one of these teaching's target application are forultra low power and portable electronics, the transistors utilized incircuits operate under the subthreshold region, but it is also possibleto operate transistors, throughout this disclosures and illustrations,in the normal regions. Throughout this disclosure and its illustrations,current sources or current mirrors may be constructed with single FETsor cascodes one, depending on cost-performance considerations such asdie size, output impedance, gain, speed, head room, amongst others.Illustrations in some embodiments utilizing NMOSFET (e.g., in a currentmirror or RGC-CM), can be modified to utilize their complementary FETcounterparts (i.e., PMOSFETs). Moreover, embodiments utilizingamplifier's with PMOSFET input stages can also be modified to utilizeNMOSFET input stages, or complementary input stage (i.e., both PMOSFETand NMOSFET operating in parallel) for rail to rail input dynamic range.Moreover, embodiments can utilize amplifiers, or their variationsobvious to one skilled in the art, that use double PMOSFETs coupled withNMOSFET level shifters or double NMOSFETs coupled with PMOSFET levelshifters to attain rail-to-rail input operations. The U.S. ProvisionalPatent Application Ser. No. 62/304,373 utilizes an amplifier's inputstage based in complementary input pairs (i.e., both PMOSFET and NMOSFEToperating in parallel) for rail to rail input dynamic range. The U.S.Provisional Patent Application Ser. No. 62/320,512 and the U.S.Provisional Patent Application Ser. No. 62/415,496 utilize anamplifier's input stage that are based in double PMOSFETs coupled withNMOSFET level shifters to attain input voltage rail-to-rail operations.For demonstrative clarity and simplicity, the assumption is made thatV_(A), β, η, g_(m), g_(ds), r_(ds), r_(O), and I_(D) for PMOSFETs andNMOSFETs are substantially equal unless otherwise specified. In order todemonstrate achievable typical specifications, simulations are performedon some of the circuits illustrating the embodiments. These simulationsare not intended to guarantee the embodiment performance to a particularrange of specifications. Circuit simulations use the Top-Spicesimulator, and are based on approximate device models for a typicalstandard 0.18 μm CMOS process fabrication. To simulate for sensitivityof design to device model variations (i.e., simulate for design margin),this disclosure provides some worst case simulations (WC) that subjectthe circuit embodiments to variations in device parameters (e.g.,manufacturing process fluctuations in V_(TH) and C_(OX)), which canindicate the circuit's design sensitivity to normal manufacturing (waferlot-to-lot) variations. For example, WC circuit simulations illustratevariations on performance specifications such as slew rate of settlingtime as a function of varying V_(TH) by ±10% from center value, andindependently from varying C_(OX) by ±5% from center value (i.e., 16process corner combinations).

Note that the teachings in this disclosure are applicable to highvoltage or high current or high speed applications, and combinationsthereof, as well. The teachings in this disclosure may be applied toother manufacturing fabrication processes that contain transistors, suchas fabrication processes with deep sub-micron CMOS, BiCMOS, BJT, andJFET, amongst other device and manufacturing platforms that make devicesthat can function as transistors.

Current mirrors, current sources, amplifier, and output buffers arebuilding blocks for any integrated circuits (IC), including for higherorder functions such as analog to digital converters (ADC), digital toanalog converters (DAC), regulators, references, fitters, dataacquisition systems (DAS), and other building blocks in any analog andmixed mode ICs, and system on a chip (SOC). It is advantageous for ICbuilding blocks such as current sources, current mirrors, amplifiers,and buffer drivers to: (1) consume low currents, (2) operate with lowpower supply voltages, especially for portable applications, (3) havenear rail-to-rail input-output spans since there is little voltageheadroom to waste, especially under low power supply voltages, (4) havehigh output impedance or high output gain, especially under ultra lowoperating currents when gain is lower, (5) have high-speeds, especiallyunder low operating currents when speed gets slower, (6) have low noise,especially under low operating currents when noise increases, (7) havetheir other performance specifications, such as power supply rejectionand common mode rejection, amongst others, unimpeded when utilizingcircuit arrangements to improve gain, speed, and noise operating underlow operating currents, and low power supply voltage conditions, (8)perform to specifications over fabrication process variations, and underdifferent power supply and temperature conditions, (9) be small so theycost less, and be able to integrate multiple building block channels onthe same die for better channel-to-channel matching, (10) use standardCMOS fabrication for lower cost, availability, and proven quality.

As the demand for portable, green electronics, and energy harvesting(e.g., self-powered ICs) continues to grow, so does the requirement forcircuits with lower power supplies and ultra low current consumptions.Low power electronics also require performance to specifications at lowV_(DD) and V_(SS) where there is less available voltage headroom toprocess signals. Meeting low power consumption and signal-to-noisespecifications at low V_(DD) and V_(SS) require ICs with the widestinput-output dynamic range possible. Although special fabricationprocesses can provide special transistors that enable a circuit tooperate with lower power supplies and lower current consumption, butsuch special manufacturing are generally expensive, and may create ahigh cost barrier for the full market potentials to be realized in atimely manner.

Also, low power (ultra low current and low power supply voltagecombined) may be required in some medical or defense applications whereit is not safe to frequently replace an implantable or embedded batteryoperated IC. This factor would require ultra low power so that the ICsdraw extremely low current so that the battery life is significantlyextended.

Another example is next generation energy harvesting electronics that iswireless and battery less. They can function perpetually without everneeding to be connected to power source and with no need of beingrecharged. There are sub-categories of energy harvesting ICs that can bedesignated as self-powered ICs. Energy harvesting is part of greenelectronics that rely on harvesting or scavenging energy from theenvironment such as solar, mechanical, thermal, or magnetic, to name afew. These kinds of energies help generate voltage potential that canfor example be stored on a super capacitor, which can power ultra lowpower electronics for signal processing.

Additionally, some biometrics system on a chip may require small diesize for applications requiring multi-channel current mirrors oramplifiers for conditioning multiple sensor's outputs simultaneously.For example, multiple sensors implanted in tooth dentures thatsimultaneously read levels of sugar, salt, acidity, temperature, andother non-vitals, which may require and at a minimum will benefit fromsmall circuit size for better matching between each circuit channels inone die.

Furthermore, to target high volume markets and reduce the risk of lowyields for long term production, manufacturing organizations generallyhave an unfavorable view of circuits that require special processes.Optimal yield and quality generally avoid circuits that requirevariations to a standard process, or complex circuits whosespecifications may depend on multiple device or (manufacturing) processparameters. Generally circuits requiring non-standard fabrication, orcomplex circuits are harder to optimize for maximal production yields,or they may compromise rugged end-product manufacturing (quality) goals.It is also of note that complex circuits may hinder transient response,in start up and turn off phases, for example. This may be a risky trait,particularly in energy harvesting applications that may subject the ICto less predictable or disorderly power supply on and off patterns inthe field.

Besides ultra low power, emerging applications such as energy harvestingthat was noted above and bio-metrics, require small circuits to keep thecosts down in order to realize their full and highest volume marketpotentials.

As stated earlier, current mirrors are fundamental building blocks inelectronics. Some of performance specifications for a current mirrorsare to have high R_(OUT) and wide input-output voltage spans. Also,making current mirrors with simple circuits that are low power and lowcost is beneficial to cost-performance-quality tradeoffs, including forhigh order other building blocks such as amplifiers, where currentmirrors are utilized.

Utilizing Regulated Cascode (RGC) is a way to increase the R_(out) of acurrent mirror. FIG. 1G is an illustration of a prior art RGC currentmirror (RGC-CM), which is not complex and yet it is low cost and rugged,but its' output-output voltage span is restrictive. As depicted in FIG.1G, the drain terminal of M_(119G) is the output of the RGC-CM, whereV_(101G)=VGS_(M119G)+VGS_(M111G). As such, the minimumV_(103G)≈Von_(M119G)+VGS_(M111G) which restricts the voltage span of thecurrent mirror's output.

Operating at low or ultra low operating currents, generally causes anamplifier's gain to be lower. Utilizing RGC-CM is a way to increase thegain of amplifiers, in part, by way of increasing the amplifier'sR_(OUT) at the amplifier's high-gain node.

Increasing an amplifier's gain, utilizing RGC-CM, without restrictingits input and output voltage span, are disclosed in the literature andin the references provided, including in [S. Yan and E.Sanchez-Sinencio, “Low voltage analog circuit design: A tutorial”, IEICETrans. Analog Integrated Circuits and Systems, vol. E83A, no. 2, pp.179-196, 2000], and [P. E. Allen & D. R. Holberg, CMOS Analog CircuitDesign, 2nd Ed, Oxford University press, 2002]. Generally, the availableRGC-CMs with a wide input-output voltage span have higher transistorcount (i.e., larger die size), consume high power, have unfavorabletransient response considering current consumption, or are complex(e.g., performance depends on different types of FETs, or performancedepends on multiple device parameters) which are generally unfavorablefor rugged manufacturing in terms of cost-performance-quality traits.

As noted earlier, low voltage electronics require rail-to-railinput-output operations. A folded cascode transconductance amplifier(FCTA), is a suitable amplifier topology for rail to rail operations.From a high level perspective, FCTA contains a common source amplifier(CSA) at its inputs. Then the differential current outputs of this CSAfeed the differential current input of a common gate amplifier (CGA)whose differential output currents are fed onto a differential input tosingle output current mirror, that sum at the FCTA high impedance outputnode to make an output voltage (V_(OUT)). The regulated cascode currentmirror (RGC-CM) is generally used in the CGA and current mirror sectionsof an amplifier to improve their R_(OUT), which increase's theamplifier's gain. Moreover, its beneficial for the RGC-CM to have wideinput-output voltage span in order not to restrict the amplifier'sinput-output voltage span, in which it is utilized. Operating the inputsof a FCTA rail to rail is generally accomplished by running pairsPMOSFET and NMOSFET (complementary) CSAs in parallel as inputs. Near therails, either PMOS CSA runs out of headroom while the NMOS CSA takescontrol and keeps feeding the next CGA gain stage, or vice versa.

Given the wide common mode range in an amplifier is especiallybeneficial for near rail-to-rail input-output voltage spans, utilizing afloating current source (FCS) or emulating its equivalent function, isgenerally a way to reduce the errors generated in upper and lowercurrent source in the CGA and current mirror (summing node) gain stage.However, in order to operate the amplifier with lowest V_(DD) (besidesoperating the amplifier inputs-outputs rail-to-rail), all of theamplifier's elements, including the FCS need to operate with lowestV_(DD). FIG. 2D is an illustration of a prior art FCS, which is lowpower and fast, but its minimum V_(DD)≥2V_(GS)+V_(IDS) would limit theamplifier for low power supply voltage applications.

Generally an amplifier's high gain stage is coupled with a bufferdriver, which makes a buffer amplifier that would be able to driveexternal loads. In order to operate the buffer amplifier with ultra lowcurrents and in low power supply environment, the buffer drivercontained in it, must also consume ultra low currents and be capable ofoperating to specification with low power supplies. As noted earlier,operating in low or ultra low currents, slows down the speed. Hence, itwould be advantageous to have a buffer driver that is fast, inherently,at low currents and can operate with low power supply voltage and havenear rail-to-rail input-output voltage spans. Running fast and operatingwith low currents at low power supplies, still requires a buffer driverwith high sink-source current drive capability to handle currentrequirements of different external loads. For example, some emergingportable applications use resistive sensors to measure environmentaltoxicity. The resistivity of such toxicity sensors can dropsignificantly when they are activated to make a measurement. Hence, itwould be advantageous to have a buffer driver that can handle externallow resistive loads when the sensor is activated, but return to lowcurrents consumptions seamlessly, when the resistive sensor is no longeractivated.

As mentioned earlier, next generation energy harvesting, and wirelessand batteryless electronics are emerging applications, that requireultra low power ICs. All else equal, operating analog ICs at ultra lowcurrents present additional challenges such as high noise besides lowgain, and slow dynamic response (e.g., as noted earlier and generally,the lower the current, the higher the noise, the lower the gain, and thelower the speed). Hence, it would be advantageous to have a low noiseamplifier noise that consumers low operating currents, and one that canbe fast and operate with low power supply voltage.

Therefore, in summary here are a list of advantages of these teachings.One, is to make current mirrors that are simple, small, low cost, lowpower, have high R_(OUT), and wider input-output voltage span, whoseembodiments are illustrated in FIG. 1B, FIG. 1D, and FIG. 1F. Two, asjust stated, low current reduce an amplifier's gain, generally. Hence,another benefit of this teaching is to improve amplifiers gain andinput-output voltage span, while keeping its power consumption low, byutilizing the disclosed RGC-CMs in amplifiers. The amplifier of FIG. 1Autilizes the RGC-CM coupled with DCSC (illustrated in FIG. 1B). Theamplifier of FIG. 1C utilizes the RGC-CM coupled with ICMA (illustratedin FIG. 1D). The amplifier of FIG. 1E utilizes the RGC-CM coupled withcomposite amplifier, CSGA (illustrated in FIG. 1F). As a result ofutilizing the disclosed RGC-CMs in the amplifiers, the amplifier'sperformance is improved, some aspects of which are illustrated insimulations FIG. 1I to FIG. 1M. Three, it would be advantageous toimprove floating current sources (FCS), or emulate the function of FCS,that can operate at low power supply voltage. By utilizing the disclosedFCSs in amplifiers, the said amplifier can operate with low V_(DD) whilemaintaining its accuracy. The embodiments of amplifiers utilizing first(FCS_(200A)), second (FCS_(200B)), and third (FCS_(200C)) FCS areillustrated in FIG. 2A, FIG. 2B, and FIG. 2C, respectively. Four,another benefit is to lower the output noise of an amplifier by narrowbanding the amplifier. To reinvigorate the speed of the narrow bandedamplifier, a boost-on signal is initiated which dynamically and rapidlyinjects a substantial current into the amplifier's bias current to speedup its slew rate, when the amplifier's inputs get off balance due to alarge transient differential input signal. Subsequently, after theamplifier has had some time to respond and regulate itself and as theamplifier's inputs approach balance, a boost-off signal dynamicallyinjects a slow and decaying (to zero) current into amplifier's biascircuitry, instead of turning the boost current rapidly, in order toimprove the amplifier's settling time. Additionally, the boost circuitryand the amplifier's time constants, that determine the dynamic responseof the boost and that of the amplifier, are approximately matched inorder to improve smooth transitions in and out of boost and to improveconsistency of the amplifier's dynamic response across process,temperature and operation variations. The embodiments of noise reductionand speed boost circuits are illustrated in FIG. 3A, and FIG. 3B.Simulations depicted in FIG. 3C and FIG. 3D illustrate some aspects ofthe improvements in noise, and the dynamic response of an amplifier withand without the disclosed noise reduction and speed boost circuits.Five, another feature of this disclosure is to make a low voltage, lowcurrent, and high-speed buffer driver that is coupled with the high gainstage of an amplifier (FIG. 4A). The buffer driver utilizes CNICM tomonitor and rectify the sink-source currents of FET that drive externalloads, before the sink-source FET current signals are fed to MCS or LTAcircuits. The output of MCS or LTA circuits is then fed onto a NICMA orICMA or CMA which helps regulate the current in the inactive sink-sourceFET. The buffer driver can operate with low power supply voltage andconsume low currents. Because the elements utilized in the buffer driverrun chiefly in current mode, it is inherently fast. Also, because theelements utilized in the buffer driver run chiefly in current mode,voltage swings are inherently small, which helps the buffer driveroperate with lower power supplies. The embodiments of buffer drivercircuits are illustrated in FIG. 4B, FIG. 4C, and FIG. 4D. Simulationsdepicted in FIG. 4E and FIG. 4F and FIG. 4G illustrate some aspects ofthe improvements in current drive capability of amplifiers (that utilizethe disclosed buffer drivers) while running at very low operatingcurrents.

Note that the following papers providing additional analysis ofrelevance to low power and low cost amplifier designs are also herebyincorporated by reference in their entirety: (1) A. Far, “Small sizeclass AB amplifier for energy harvesting with ultra low power, highgain, and high CMRR,” 2016 IEEE International Autumn Meeting on Power,Electronics and Computing (ROPEC), Ixtapa, Zihuatanejo, Mexico, 2016,pp. 1-5; (2) A. Far, “Amplifier for energy harvesting: Low voltage,ultra low current, rail-to-rail input-output, high speed,” 2016 IEEEInternational Autumn Meeting on Power, Electronics and Computing(ROPEC), Ixtapa, Zihuatanejo, Mexico, 2016, pp. 1-6; (3) A. Far, “LowNoise Rail-To-Rail Amplifier for Energy Harvesting Runs Fast at UltraLow Currents,” 2017 IEEE Canadian Conference on Electrical and ComputerEngineering (CCECE), Windsor, ON, 2017; and (4) A. Far, “Ultra LowCurrent and Low Voltage Class AB Buffer Amplifier,” 2017 IEEE CanadianConference on Electrical and Computer Engineering (CCECE), Windsor, ON,2017.

Section (I): Detailed Description of Regulated Cascode Current Mirror(RGC-CM) Coupled with Diode Connected Self Cascode (DCSC), asIllustrated in FIG. 1B

FIG. 1B is a circuit schematic showing a RGC-CM_(100B) coupled withDCSC_(100B) according to an embodiment. One differentiation of thisteaching, compared to the conventional teachings, is in arranging aRGC-CM_(100B) coupled with DCSC_(100B) to generate a DC voltage shiftthereby allowing the input-output voltage of the current mirror to getcloser to the rail.

The basic idea of RGC-CM_(100B) coupled with DCSC_(100B) in FIG. 1B, isto use negative feedback of an auxiliary amplifier (A_(AUX)) that has again of ‘G’ (e.g., formed by M_(111B) and I_(105B)), which regulates theV_(DS) of M_(117B) as constant as possible, irrespective of thevariations in the output node 102B of the current mirror, which is V_(D)of M_(119B). Here, the DCSC shifts down the V_(DS) of M_(117B) down byVDS_(M113B) which helps V_(102B) (output of current mirror) get closerto V_(SS).

Throughout the description of FIG. 1B, ‘X’ is the W/L of a MOSFET, and‘r’, ‘s’, ‘t’, ‘u’, ‘v’, and ‘w’ are scale factors for MOSTEL W/Ls.These MOSFET scale factors can be set approximately in the ranges of0.01≤r≤100, 0.01≤s≤100, 0.01≤t≤100, 0.01≤u≤100, 0.01≤v≤100, and0.01≤w≤100 depending on considerations such as current consumption,voltage span, and die size, amongst others. Also, current source (e.g.,I_(111B), I_(105B), and I_(107B)) scale factors can be set approximatelyin the ranges of 0.01≤o≤100, 0.01≤p≤100, 0.01≤q≤100. For example, for aembodiment of FIG. 1A that utilizes plurality of RGC-CM_(100B) coupledwith DCSC_(100B) of FIG. 1B, the MOSFET's W=4 μm, and X=1 μm computes toW/L=4. In this example, the MOSFET W/L scale factors are set to r=10,v=3, and u=t=s=t=w=1. Also, here the current sources, i=5 nA forI_(M111B)=I_(M105B)=I_(M107B)=5 nA, where o=p=q=1.

The connections of the elements in FIG. 1B are described as follows. Thebody terminal of all NMOSFETs in FIG. 1B are connected to node 2 whichis V_(SS), and the body terminals of PMOSFETs are connected to node 1which is V_(DD). Bias current sources I_(111B), I_(105B), and I_(107B)have their upper terminals connected to node 1, which is V_(DD). Thesource terminal of NMOSFETs M_(126B), M_(111B), and M_(117B) areconnected to node 2 that is V_(SS). The lower terminal of I_(111B) isconnected to node 100B, which is connected to the gate terminal ofM_(126B), and the drain terminal of M_(126B), and the gate terminal ofM_(117B). Node 101B is connected to the lower terminal of I_(105B), thedrain terminal of M_(111B), and the gate terminal of M_(119B). Node 102Bis connected to the drain terminal of M_(119B), which is the output ofthe current mirror (or terminal Io_(100B)). Node 103B is connected tothe gate terminal of M_(111B), the source terminal of M_(115B), and thedrain terminal of M_(113B). Node 104B is connected to the lower terminalof I_(107B), the gate terminal of M_(115B), the drain terminal ofM_(115B), and the gate terminal of M_(113B). Node 105B is connected tothe source terminal of M_(113B), the drain terminal of M_(117B), and thesource terminal of M_(119B). The I_(M111B) terminal is the input currentto the RGC-CM_(100B) coupled with DCSC_(100B), and I_(M119B) terminal isits output.

Describing the details of the circuit in FIG. 1B is as follows. TheI_(111B) flows through M_(126B), whose V_(GS) is the same as that ofM_(117B), which mirrors and scales v×I_(M126B)=u×I_(M117B). Fordescriptive clarity of FIG. 1B, let's assume current source scalefactors of o=q=1, and the MOSFET scale factors of r=10, s=t=u=w=1, andv=2. As such, I_(107B)≈1i (which flows through M_(115B) and M_(113B)) isfed into node 105B where ID_(M117B)≈2i, and hence the current mirror'soutput which is ID_(M119B)≈ID_(M117B)−I_(107B)≈1i.

The current mirror of FIG. 1B utilizes DCSC which is composed of twoFETs, M_(115B) and M_(113B), whose function is briefly discussed. Thebasic function of DCSC here is to provide a DC voltage shift, which is afunction of the difference(ΔVGS)=VGS_(M113B)−VGS_(M115B)≈VDS_(M113B)≈V_(T)×ln(r/s). In thesubthreshold region, this DC voltage shift is chiefly dependent on V_(T)which is well controlled and it is approximately independent ofCMOSFET's V_(TH) (note that the value of V_(TH) has normal but widefluctuations in fabrication manufacturing).

In FIG. 1B, the A_(AUX) which is configured as a simple common sourceamplifier (CSA) which can be composed of only one FET and one currentsource, M_(111B)-I_(105B), where I_(105B) can be supplied via a one FETof a PMOS current mirror. This A_(AUX), has an approximate gain ofG∝V_(A)/V_(T), and it regulates the gate voltage of M_(119A) in order toincrease the R_(OUT) at node 102B, which is the output of the RGC-CM.The R_(OUT) is approximately ∝g_(m) ²×r_(ds)³∝(V_(A)/V_(T))²×(V_(A)/I_(D)). Note that R_(OUT) of RGC-CM on actualsilicon will likely be lower due to second order effects such as η(subthreshold slope factor) and substrate leakage, amongst otherfactors. Moreover, it is of note that for additional headroom atRGC-CM's input-output, multiple DCSCs can be cascaded to developed alarger DC voltage shift, for example.

In summary, the prior art illustrated in FIG. 1G, the minimumV_(103G)=Von_(M119G)+VGS_(M111G). Comparatively, the embodiment of FIG.1B utilizes a simple circuit that improves the minimumV_(102B)≈Von_(M119B)+VGS_(M111B)−V_(T)×ln(r/s). Also note thatVDS_(M117B)=VGS_(M103B)−VDS_(M113B), andVGS_(M115B)−VGS_(M113B)+VDS_(M117B)≈VDS_(M111B)≈V_(T)×ln(r/s).

In conclusion, the benefits of the FIG. 1B, RGC-CM_(100B) coupled withDCSC_(100B) include the following. First, the same type FETs may be usedin each of the A_(AUX) (e.g., a M_(111B) which is NMOSFETs), theDCSC_(100B) (e.g., M_(115B)-M_(113B) which are also NMOSFETs), and thecascode FETs in the current mirror (e.g., M_(117B)-M_(119B)) whosedevice parameters match and track, and hence benefiting the amplifier'sDC, AC, and transient specifications stability over fabrication process,temperature, and power supply variations. Second, the bias voltages inthe RGC-CM_(100B) coupled with DCSC_(100B) are self biased (i.e., noneed for separate voltages to set its internal bias nodes), which lowercomplexity and lowers current consumption, reduces die size and cost,and improved quality. Third, the RGC-CM_(100B) coupled with DCSC_(100B)has low transistor count (i.e., it is small) and low current becauseits' auxiliary amplifier's gain is achieved by only two FETs and thewidening of output voltage headroom is achieved by only two FETs (e.g.,M_(115B) and I_(113B)) in DCSC. Fourth, the RGC-CM_(100B) coupled withDCSC_(100B)'s headroom is widened by V_(T)×ln(r/s) with 2 FETs, whichcan practically be programmed or pre-determined to have a range of, forexample, 50 mV to 150 mV. For sub-i V power supply environments, every100 mV extra headroom in the current mirror translates to 10% extraoperating room at the current mirror output. Fifth, as stated earlier,electronic functions used in energy harvesting applications may requiremore smooth and stable power up and down transient responses (e.g., forpower save reasons or for frequent switching from a magnetic powerharvester source to a kinetic power harvester source). The simple andsmall circuit arrangement, in RGC-CM_(100B) coupled with DCSC_(100B)that utilizes few FETs (i.e., 2 FETs to make a single stage A_(AUX) toincrease R_(out)), is beneficial for AC and transient response of thecurrent mirror. This is because the single stage amplifier neitherrequire multiple AC loops nor complicated AC and transient compensation.For energy harvesting applications, that may subject ICs to lesspredictable or disorderly power supply on and off patterns, this simplearrangement in RGC-CM coupled with DCSC can provide additional guardband for smoother dynamic performance.

Section (II): Detailed Description of Regulated Cascode Current Mirror(RGC-CM) Coupled with Inverting Current Mirror Amplifier (ICMA), asIllustrated in FIG. 1D

FIG. 1D is a circuit schematic showing a RGC-CM_(100D) coupled withICMA_(100D), according to another embodiment. A differentiation of thisteaching, compared to conventional teachings, is in arranging aRGC-CM_(100D) coupled with ICMA_(100D), where ICMA_(100D) has a currentinput terminal whose operating voltage can be programmed withflexibility. As a result, this aspect of ICMA_(100D) can widen theinput-output voltage range of the current mirror. Here, the ICMA_(100D)serves the function of auxiliary amplifier, A_(AUX), that increase itsR_(OUT). The benefits of this arrangement include the widening ofvoltage output span, and increasing the R_(OUT) of the RGC-CM_(100D)coupled with ICMA_(100D) can provide low power and high speed with lowcost. Throughout the description of FIG. 1D, ‘X’ is the W/L of a MOSFET,and ‘n’, ‘r’, ‘s’, ‘t’, ‘u’, ‘v’, and ‘w’ are scale factors for MOSTELW/Ls. These MOSFET scale factors can be set approximately in the rangesof 0.01≤n≤100, 0.01≤r≤100, 0.01≤s≤100, 0.01≤t≤100, 0.01≤u≤100, 0.01≤v100, 0.01≤w≤100 depending on considerations such as current consumption,voltage span, and die size, amongst others. Also, for current source(e.g., I_(113D), I_(105D), I_(107D), and I_(109D)) scale factors can beset approximately in the ranges of 0.01≤o≤100, 0.01≤p≤100, 0.01≤q≤100,and 0.01≤m≤100. For example, for an embodiment of FIG. 1C that utilizesRGC-CM_(100D) coupled with ICMA_(100D) of FIG. 1D, for MOSFET's W=4 μm,and X=1 μm, which computes to W/L=4. In this example, the MOSFET W/Lscale factors are set to n=0.05, v=3, and r=s=t=u=w=1. Also, currentsource scale factors can be set to q=2, and o=p=m=1 for i=5 nA such that

$I_{M\; 111D} \approx I_{M\; 113\; D} \approx I_{M\; 115D} \approx I_{M\; 121D} \approx I_{M\; 119D} \approx {\frac{1}{2} \times I_{M\; 117\; D}} \approx I_{M\; 128\; D} \approx {5\mspace{14mu}{{nA}.}}$

The connections of the elements in FIG. 1D are described as follows. Thebody terminal of all NMOSFETs in FIG. 1D are connected to V_(SS), andthe body terminals of PMOSFETs are connected to node 1 that is V_(DD).Bias current sources I_(113D), I_(105D), I_(107D), and I_(M109D) havetheir upper terminals connected to node 1 that is V_(DD). The bodyterminal of all NMOSFETs in FIG. 1B are connected to node 2 that isV_(SS), and the body terminals of PMOSFETs are connected to node 1 thatis V_(DD). The source terminal of NMOSFETs M_(128D), M_(111D), M_(113D),M_(117D), and M_(121D) are connected to node 1 that is V_(SS). The lowerterminal of I_(113D) is connected to node 100D, which is connected tothe gate terminal of M_(128D), and the drain terminal of M_(128D), andthe gate terminal of M_(117D). Node 101D is connected to the lowerterminal of I_(105D), the drain terminal of M_(111D), and the gateterminal of M_(119D). Node 102D is connected to the drain terminal ofM_(115D), the drain terminal of M_(113D), the gate terminal of M_(113D),and the lower terminal of I_(107D). Node 103D is connected to the drainterminal of M_(119D), which is the output of the current mirror (orterminal Io_(100D)). Node 104D is connected to the source terminal ofM_(115D), the drain terminal of M_(117D), and the source terminal ofM_(119D). Node 105D are connected to the gate terminal of M_(121D), thedrain terminal of M_(121D), and the lower terminal of I_(109D). TheI_(M113D) is the input current to the RGC-CM_(100D) coupled withICMA_(100D), and ID_(M119D) is its output current that is its mirror.

Note that the ICMA_(100D) function is performed by M_(115D), M_(113D),I_(107D), M_(111D), and I_(105D). For clarity in describing theRGC-CM_(100D) coupled with ICMA_(100D) of FIG. 1D, it is assumed thatn=0.05, v=2, and r=s=t=u=w=1. Also, it is assumed that the currentsource scale factors to q=2, and o=p=m=1 for i=5 nA.

During steady state conditions, I_(113D) flows through M_(128D), whoseV_(GS) is the same as that of M_(117D), which causesv×I_(M128D)=u×I_(M117D) or 2I_(M128D)=I_(M117D). As such,ID_(M111D)≈ID_(M113D)≈ID_(M115D)≈ID_(M121D)≈ID_(M119D)≈½×ID_(M117D)≈ID_(M128D)≈i≈5nA.

As noted earlier, the regulated cascode (RGC) is used in a currentmirror (CM), chiefly, to increase its R_(OUT). When, there are voltagechanges (Δv_(X)) at the output (node 103D), the goal is for the outputcurrent (ID_(M119D)=i) variations (Δi_(X)) to be minimized. The Δv_(X)here can cause a Δi_(X) on M_(119D), because M_(119D) has a finiteimpedance between its source and drain terminals. SinceI_(M117D)=2I_(128D)=2i is fixed, then at node 104A, the Δi_(X) wouldflow through M_(115D), which is the current input terminal ofICMA_(100D). Given that I_(107D)=2i is also constant, as explainedearlier, this change in current, Δi_(X), flowing through M_(115D) getssubtracted from the steady state current of ID_(M113D). The auxiliaryamplification, A_(AUX), function inside the ICMA_(100D) is performed byM_(111D) and I_(105D) whose R_(OUT)∝V_(A)/I_(D) (when CMOS is operatingin the subthreshold region). Consequently, the Δi_(X) in ID_(M113D) ismirrored onto M_(111D) whose drain (at node 101D is coupled toI_(105D)=i that is a constant current source) responds to this Δi_(X),with negative gained voltage change of about Δi_(X)×R_(OUT). At node101D, this negative gained voltage change is fed-back to the gateterminal of M_(119D). This negative gain voltage feedback, is themechanism that regulates the VGS of M_(119D) so that Δi_(X)→near zero.

In summary, the ICMA_(100D) regulates the gate voltage of M_(119D) inorder to increase the R_(OUT) of at node 103D. The R_(OUT) ofRGC-CM_(100D) coupled with ICMA_(100D) is approximately ∝g_(m) ²×r_(ds)³∝(V_(A)/V_(T))²×(V_(A)/I_(D)), assuming subthreshold operations andassuming that the gain from node 104D to 102D is about unity with equalcurrents flowing through M_(115D), M_(113D), and M_(111D).

The RGC-CM_(100D) coupled with ICMA_(100D) output voltage span is alsoimproved substantially, mainly because of the flexibility in settingV_(105D) fairly independently (e.g., V_(105D)∝I_(109D), and W/L ofM_(121D)) which bias the VG_(M115D). The minimumV_(103D)=Von_(M119D)+Von_(M117D), andVon_(M117D)=VGS_(M121D)−VGS_(M115D)=ΔVGS≈V_(T)×ln(r/n). Hence, minimumV_(103D)=Von_(M119E)+V_(T)×ln(r/n) above V_(SS), and approximatelyindependent of CMOSFET's V_(TH).

In conclusion, the benefits of the proposed RGC-CM_(100D) coupled withICMA_(100D) illustrated in of FIG. 1D include the following. First itcontains the function of the auxiliary amplifier that is composed of thesame type FETs (e.g., a M_(115D), M_(113D), and M_(111D) that are allNMOS) as that of the FETs in the cascoded current mirror (e.g.,M_(117D), M_(119D) that also NMOS). Note that constant current sourcesID_(105D) and ID_(107D) can be made of single PMOSFETs to serve only thefunction of constant ‘bias current’ source. Therefore, the DC, AC, andtransient specifications stability and manufacturability of RGC-CM canbe improved, because utilizing the device parameters of the same type ofFETs (e.g., NMOSFETs) better match and track each other over normalvariations in fabrication process, temperature, and power supply.Second, RGC-CM_(100D) coupled with ICMA_(100D) has low transistor count(i.e., it is small) and low current because the expanded output span aswell as the auxiliary amplification function to increase R_(OUT) can beaccomplished by 4 FETs (M_(111D), M_(113D), M_(115D), M_(121D)) and 3current sources (I_(105D), I_(107D), and I_(109D) which can be made with3 FETs as well). Third, it can operate fast in part because ICMA operatein current mode. Moreover, ICMA is fast because small geometry FETs canbe selected in the ICMA signal path (M_(111D), M_(113D), and M_(115D))with a scale factor of 1 (i.e., they are not composed of plurality ofFETs arranged in parallel). Fourth, the headroom is widenedsubstantially allowing the RGC-CM_(100D) coupled with ICMA_(100D)'soutput voltage to swing within Von_(M119E)+V_(T)×ln(r/n) above the powersupply rail. As noted earlier, for sub-1V power supply environments,every 100 mV extra headroom in the current mirror translates to 10%extra operating room at the current mirror output. Fifth, as statedearlier, electronic functions used in energy harvesting applications maysubject ICs to less predictable or disorderly power supply on and offpatterns, thereby requiring more smooth transient responses. ThisICMA_(100D) operates chiefly in current mode which is inherently fast,and its A_(AUX) has a single pole (simplifying its compensation) thatcan provides for additional transient performance guard band.

Section (III): Detailed Description of Regulated Cascode Current Mirror(RGC-CM) Coupled with Composite Amplifier (CSGA), Illustrated in FIG. 1F

FIG. 1F is a circuit schematic showing a RGC-CM_(100F) coupled withCSGA_(100F) according to a embodiment. As explained in previoussections, increasing the R_(OUT) of a current mirror is one of themotivation for utilizing an auxiliary amplifier (A_(AUX)). One of thedifferentiation of the RGC-CM_(100F) coupled with CSGA_(100F) in FIG. 1Fis in the arrangement of CSGA_(100F) that performs the function ofA_(AUX) to increase R_(o) as well as input-output voltage span of thecurrent mirror by utilizing a few FETs. Here, the CSGA_(100F) utilizes afirst differential common source amplifier (CSA), with a built-inoffset, that feeds its output current into first a common gate amplifier(CGA) that is inherently fast and whose gate is biased from the commonsource summing junction of the first CSA. The designation of ‘compositeamplifier’ is because this CSGA_(100F) intertwines the composite ofPMOSFETs and NMOSFETs to generate both the gain and widen the input-outvoltage span of the current mirror. The gain ‘G” (attained chiefly byM_(113F) and I_(109F)) of CSGA_(100F) keeps the V_(DS) of M_(117F) asconstant as possible, ideally speaking, irrespective of the variationsin V_(D) of M_(119F). Here, the input-output voltage span of the currentmirror is widened by lowering the VDS_(M117F). This voltage is set bythe built-in input offset voltage (ΔVGS VDS_(M117F)≈V_(T)×ln(r/s))generated by scaling M_(115F)-M_(111F) that are the differential inputsthe first CSA within the CSGA_(100F). Alternatively, (to provide the CSAwith a DC voltage shift), the M_(111F) input of CSA can be biased at analternative voltage instead of ground or V_(SS), which avoids scalingM_(115F)-M_(111F) and saves die area.

Throughout the description of FIG. 1F, ‘X’ is the W/L of a MOSFET, and‘r’, ‘s’, ‘t’, ‘u’, ‘v’, and ‘w’ are scale factors for MOSTEL W/Ls.These MOSFET scale factors can be set approximately in the ranges of0.01≤r≤100, 0.01≤s≤100, 0.01≤t≤100, 0.01≤u≤100, 0.01≤v≤100, and0.01≤w≤100 depending on considerations such as current consumption,voltage span, and die size, amongst others. Also, current sources (e.g.,I_(111F), I_(107F), I_(109F), and I_(119F)) scale factors can be setapproximately in the ranges of 0.01≤n≤100, 0.01≤o≤100, 0.01≤p≤100,0.01≤q≤100. For example, for an embodiment of FIG. 1E that utilizesRGC-CM coupled with composite amplifier of FIG. 1F, the MOSFET's W=4 μm,and X=1 μm computes to W/L=4. In this example, the MOSFET W/L scalefactors are set to s=0.0625, v=2, and u=t=r=t=w=1. Also, here thecurrent sources, i=5 nA for2×I_(M119F)=2×I_(M109F)=I_(M111F)=I_(M107F)=10 nA, where q=o=1, n=p=2.

The connections of the elements in FIG. 1F are described as follows. Thebody terminal of all NMOSFETs in FIG. 1F are connected to node 2 whichis V_(SS), and the body terminals of PMOSFETs are connected to node 1which is V_(DD). Bias current sources I_(119F), I_(107F), and I_(109F)have their upper terminals connected to node 1, which is V_(DD). Thelower terminal of I_(111F) is connected to node 2, which is V_(SS). Thedrain terminal of M_(111F), and the gate terminal of M_(111F) areconnected to node 2, which is V_(SS). The source terminal of NMOSFETsM_(131F) and M_(117F) are connected to node 2 that is V_(SS). Node 100Fis connected to drain terminal of M_(131F), the gate terminal ofM_(131F), and the gate terminal of M_(117F). Node 101F is connected tosource terminal of M_(111F), the source terminal of M_(115F), the gateterminal of M_(113F), and the lower terminal of current source I_(107F).Node 101F is the summing junction of CSA containing the CSA differentialinput FETs (M_(111F), M_(115F)) and the current source I_(107F) thatprovides the tail current to the CSA differential input FETs (M_(111F),M_(115F)). Node 102F is connected to drain terminal of M_(113F), and thegate terminal of M_(119F), and the lower terminal of I_(109F). Node 103Fis connected to source terminal of M_(113F), the drain terminal ofM_(115F), and the upper terminal of I_(111F). Node 104F is connected tosource terminal of M_(119F), the drain terminal of M_(117F), and thegate terminal of M_(115F). Node 105F is connected to the drain terminalof M_(119F) and it is the output terminal, IO_(100F), of the RGC-CM.

The details of various functions of the circuit in FIG. 1F, which isRGC-CM_(100F) coupled with CSGA_(100F) is now described. The I_(119F)flows through M_(131F), whose V_(GS) is the same as that of M_(117F),which mirrors and scales v×ID_(M131F)=u×ID_(M117F). For descriptiveclarity, it is assumed that current source scale factors of q=o=1,n=p=2, and the MOSFET scale factors of r=0.0625, s=t=u=w=1, and v=1. Assuch, ID_(M109F)≈1i flows through M_(M113F), which subtracts fromI_(111F)≈2i thus leaving ID_(M115F)≈1i. Consequently, ID_(M115F)≈1isubtracts from I_(107F)≈2i which leaves ID_(M111F)≈1i.

The CSGA_(100F) of FIG. 1F serves at least two functions, which is firstto provide a build-in offset (ΔVGS) that widens the input-output voltagespan of the current mirror. The ΔVGSVGS_(M115F)−VGS_(M111F)=VDS_(M117F)≈V_(T)×ln(r/s). Note that thebuild-in offset is chiefly dependent on V_(T) (generally, constant valueat ˜26 mV by laws of device physics) and approximately independent ofCMOSFET's V_(TH) (note that the value of V_(TH) has normal but widefluctuations in fabrication manufacturing). The second function of theCSGA_(100F) is to serve as the A_(AUX) as follows. A change in voltageat node 105F which is the output of the current mirror, ΔV_(OUT),generate a differential current, Δi_(OUT), at node 104F. SinceID_(M117F) is constant, then Δi_(OUT) causes a voltage change at node104F, Δv_(X). This Δv_(X) is effectively applied (to a CSA with itsinputs as M_(115F)-M_(111F)) between the nodes 104F (VDS_(M117F)) andV_(SS), setting aside the built in offset voltage which is also aconstant voltage≈V_(T)×ln(r/s). With Δv_(X) applied to CSA, an inversecurrent output change, −Δi_(X) is generated that is fed into a CGA(containing M_(113F) and I_(M109F)) that is inherently fast. The −Δi_(X)applied to the output of this CGA (at its high impedance node) generatesa negative gained voltage change at node 102F that regulates theVG_(M119F). It can be noted in FIG. 1F that while ΔVGS can be generatedby scaling PMOSFETs, but ΔVGS is independent of PMOSFET deviceparameters and it is approximate a function of V_(T) (thermal voltage).

As indicated earlier, it is possible to connect the gate terminal ofM_(111F) (the other input of the CSA) to a secondary bias voltageinstead of V_(SS) or GND that established the VG_(M111F) (e.g.,VG_(M111F)+VGS_(M111F)−VGS_(M115F)≈VDS_(M117F) V_(DS)), which would thenallow to make r=s=1. This arrangement would save area since M_(115F) orM_(111F) needs not be scaled, and the secondary bias voltage wouldfacilitate the headroom needed at node 104F

Node 101F which is the summing junction of CSA (source terminals ofM_(111F) and M_(115F)) establishes a DC voltage that biases VG_(M113F)which is the gate terminal of the CGA. The CGAs are inherently fast andCSA output (drain terminal of M_(115F)) is fed into the current input ofCGA (source terminal of M_(113F)). Therefore, the manner of arrangingthe RGC-CM_(100F) coupled with CSGA_(100F) in FIG. 1F serves thefunction of auxiliary amplifier and can speed up its dynamic response.

In summary, voltage movements, Δv_(OUT), at node 105F (output terminalof the current mirror) cause a voltage, Δv_(X), at node 104F. The Δv_(X)would cause a negative change in current, −Δi_(X), in node 103F, whichin turn cause a gained voltage change at the output of the CSGA_(100F),≈−Δi_(X)×−R_(OUT aux)≈−G×Δv_(X) at node 102F, which is the gate ofM_(119F). As a result, the source voltage of M_(119F) follows its gatevoltage, thereby regulating the voltage at node 104F until Δv_(X)→nearzero again and as such the R_(OUT) of the RGC-CM can be increased inthis arrangement.

The gain, ‘G’, of the RGC-CM amplifier is approximately G∝V_(A)/V_(T),since FETs operate in the subthreshold region. The R_(out) of RGC-CM isapproximately ∝g_(m) ²×r_(ds) ³∝(V_(A)/V_(T))²×(V_(A)/I_(D)). As statedearlier, the R_(OUT) of RGC-CM on actual silicon will be lower due tosecond order effects such as η (subthreshold slope factor) and substrateleakage, amongst other factors.

In summary, the benefits of the FIG. 1F RGC-CM₁₀₀₀ coupled withCSGA_(100F) include the following. First, same channel type FETs areutilized in the current mirrors (M_(117F), M_(119F)) and CGA (containingM_(113F)), whose device parameters match and track, and hence benefitingthe amplifier's DC, AC, and transient specifications stability overfabrication process, temperature, and power supply variations. Thegenerated built-in offset is made using PMOSFETs where the offsetvoltage is chiefly a function V_(T), and it is independent of PMOSFETdevice parameters (because they operate in the subthreshold region).Second, the bias voltages in the RGC-CM_(100F) coupled with CSGA_(100F)are self biased (i.e., no need for separate voltages to set its internalbias nodes), which lowers complexity and current consumption. Third, ithas low transistor count (i.e., it is small) and low current becauseits' auxiliary amplifier's gain function and the widening ofinput-output voltage span is achieved by 3 FETS (M_(111F), M_(113F), andM_(115F)) and 3 current sources (I_(103F), I_(107F), and I_(109F)) whereeach current source can be made of a single FET. Fourth, headroom to therails can be set (e.g., V_(T)×ln(r/s) above V_(SS)) independent of theauxiliary amplifier, and practically be programmed or pre-determined tohave a range of, for example, 50 mV to 150 mV above V_(SS). Therefore,the input-output voltage span of the current mirror is widened. As notedearlier, for sub-1V power supply environments, every 100 mV extraheadroom in the current mirror translates to 10% extra operating room atthe current mirror output.

Section (IV): Detailed Description of Amplifier (AMP_(100A)) isIllustrated in FIG. 1A, Utilizing Plurality of RGC-CM_(100B) Coupledwith DCSC_(100B) as Illustrated in FIG. 1B

FIG. 1A is a circuit schematic showing a embodiment of a folded cascodetransconductance amplifier (FCTA) that utilizes the plurality ofRGC-CM_(100B) coupled with DCSC_(100B) illustrated in FIG. 1B. When theRGC-CM_(100B) coupled with DCSC_(100B) is utilized in the FCTA, itexpands the amplifier's input-output voltage span and increases itsoutput impedance (R_(OUT)) which in turn increases the amplifier's gain(A_(V)). Beside higher gain and wider input-output voltage span, theother improvements associated with RGC-CM_(100B) coupled withDCSC_(100B), are carried over to the amplifier which improve heamplifier's gain, enables its input and output to operate rail-to-rail,lower its cost, lower its power, and increase its speed.

Note that alternative amplifier embodiments are possible such as anamplifier with NMOS input stage or complementary (PMOS and NMOS)rail-to-rail input stages, other amplifier topologies that are not FCTA,amongst others. The connections of the elements of AMP_(100A) of FIG. 1Aare described as follows. The body terminal of all NMOSFETs in FIG. 1Aare connected to node 2 that is the V_(SS), and the body terminals ofPMOSFETs are connected to node 1 that is the V_(DD). The upper terminalsof the bias current sources I_(108A), I_(109A), I_(110A), I_(111A),I_(101A), I_(103A), I_(105A), and I_(M107A) are connected to node 1 thatis V_(DD). Bias current sources I_(100A), I_(102A), I_(104A), andI_(106A) have their lower terminals connected to node 2 that is V_(SS).The source terminal of PMOSFETs M_(100A), M_(106A), M_(111A), andM_(116A) are connected to node 1 which is the positive supply voltage,V_(DD). The source terminal of NMOSFETs M_(122A), M_(126A), M_(101A),M_(107A), M_(111A), and M_(117A) are connected to the negative supplyvoltage, V_(SS). Node 100A is connected to the gate terminal ofM_(120A). Node 101A is connected to the gate terminal of M_(121A). Node100A is the V_(IN+) terminal of the amplifier and node 101A is theV_(IN−) terminal of the amplifier. Node 102 is connected to the sourceterminal of M_(120A), the source terminal of M_(121A), and the lowerterminal of I_(108A). Node 103A is connected to the drain terminal ofM_(122A), the gate terminal of M_(125A), and the lower terminal ofI_(109A). Node 104A is connected to the drain terminal of M_(123A), thegate terminal of M_(123A), the gate terminal of M_(124A), and the lowerterminal of I_(110A). Node 105A is connected the drain terminal ofM_(124A), the source terminal of M_(123A), and the gate terminal ofM_(122A). Node 106A is connected to the drain terminal of M_(126A), thesource terminal of M_(125A), and the source terminal of M_(124A). Node107A is connected to the drain terminal of M_(125A), the lower terminalof I_(111A), the gate terminal of M_(126A), the gate terminal ofM_(107A), and the gate terminal of M_(117A). Node 108A is connected tothe drain terminal of M_(107A), the source terminal of M_(109A), and thesource terminal of M_(103A) as well as the drain terminal of M_(120A).Node 109A is connected to the drain terminal of M_(117A), the sourceterminal of M_(119A), and the source terminal of M_(113A) as well as thedrain terminal of M_(121A). Node 110A is connected to the drain terminalof M_(101A), the gate terminal of M_(109A), and the lower terminal ofI_(101A). Node 111A is connected to the drain terminal of M_(105A), thegate terminal of M_(105A), the gate terminal of M_(103A), and the lowerterminal of I_(103A). Node 112A is connected the drain terminal ofM_(103A), the source terminal of M_(105A), and the gate terminal ofM_(101A). Node 113A is connected to the drain terminal of M_(108A), thegate terminal of M_(106A), the gate terminal of M_(116A), and the drainterminal of M_(109A). Node 114A is connected to the drain terminal ofM_(111A), the gate terminal of M_(119A), and the lower terminal ofI_(105A). Node 115A is connected to the drain terminal of M_(115A), thegate terminal of M_(115A), the gate terminal of M_(113A), and the lowerterminal of I_(107A). Node 116A is connected the drain terminal ofM_(113A), the source terminal of M_(115A), and the gate terminal ofM_(111A). Node 117A is connected the drain terminal of M_(102A), thesource terminal of M_(104A), and the gate terminal of M_(100A). Node118A is connected to the drain terminal of M_(100A), the gate terminalof M_(108A), and the upper terminal of I_(100A). Node 119A is connectedto the drain terminal of M_(106A), the source terminal of M_(108A), andthe source terminal of M_(102A). Node 120A is connected to the drainterminal of M_(104A), the gate terminal of M_(104A), the gate terminalof M_(102A), and the upper terminal of I_(102A). Node 121A is connectedto the drain terminal of M_(116A), the source terminal of M_(118A), andthe source terminal of M_(112A). Node 122A is connected to the drainterminal of M_(110A), the gate terminal of M_(118A), and the upperterminal of I_(104A). Node 123A is connected to the drain terminal ofM_(114A), the gate terminal of M_(114A), the gate terminal of M_(114A),and the upper terminal of I_(106A). Node 124A is connected the drainterminal of M_(112A), the source terminal of M_(114A), and the gateterminal of M_(110A). The high impedance (high gain) output ofAMP_(100A), is V_(OUT), which is node 125A that is connected to thedrain terminal of M_(118A), and the drain terminal of M_(119A).

Note that five RGC-CM_(100B) coupled with DCSC_(100B) are utilized inthe amplifier, AMP_(100A), embodiment of FIG. 1A. The NMOS ones areRGC_(101A) containingM_(122A)-M_(123A)-M_(124A)-M_(125A)-M_(126A)-I_(109A)-I_(110A);RGC_(102A) containingM_(101A)-M_(103A)-M_(105A)-M_(107A)-M_(109A)-I_(101A)-I_(103A); andRGC_(103A) containingM_(111A)-M_(113A)-M_(115A)-M_(117A)-M_(119A)-I_(105A)-I_(107A). The PMOSones are RGC_(104A) containingM_(100A)-M_(102A)-M_(104A)-M_(106A)-M_(108A)-I_(100A)-I_(102A); andRGC_(105A) containingM_(110A)-M_(112A)-M_(114A)-M_(116A)-M_(118A)-I_(104A)-I_(106A). It wouldbe possible that AMP_(100A) could function properly without RGC_(101A)and RGC_(104A), which can be substituted with diode connected cascodesfor current mirrors. The RGC_(101A) and RGC_(104A) are added in theembodiment of FIG. 1A, in part, for better systematic matchingconsiderations and improved AC performance. Note that, in order to avoidrepeating how the amplifier is benefited by each of RGC_(101A) toRGC_(105A), the explanation of one RGC-CM (e.g., RGC_(103A)) is deemedsufficient, regarding the function of RGC-CM (coupled with DCSC) in themain amplifier, FCTA.

For explanation regarding RGC-CM_(100B) coupled with DCSC_(100B) referto its detailed description, but below is a brief description of howthey operate and benefit a FCTA. The FCTA has generally 3 parts, ‘commonsource amplifier’ (CSA), ‘common gate amplifier’ (CGA), and a currentmirror (CM). The V_(IN) is applied to a differential CSA, containingM_(120A) and M_(121A), whose output feed the CGA, containing M_(109A)and M_(119A). The differential outputs of this CGA feed the CM, made upof M_(106A) and M_(116A), to make a single ended output, V_(OUT) whichis also the high impedance node (125A) of FCTA. The current I_(111A)=icontrols VGS_(M126A) that establishes I_(M126A), which is mirrored andscaled onto I_(M107A)=M_(117A)=3i. The CSA's input currentsI_(M120A)M_(121A)≈i are fed into the source terminals of M_(109A) andM_(119A), respectively, which are the differential inputs of the CGAs.The I_(103A)=i and I_(105A)=i flow through the DCSCs of RGC_(102A) andRGC_(103A), respectively, which are passed onto M_(107A) and M_(117A),and in that order. Therefore, the operating currents of CGA's(containing M_(109A) and M_(119A)) areM_(109A)≈I_(M107A)−I_(103A)−I_(M120A)≈3i−2i=i andI_(M119A)≈I_(M117A)−I_(105A)−I_(M121A)≈3i−2i=i. As explained in theRGC-CM coupled with DCSC in previous section regarding RGC_(103A), theauxiliary amplifier (composed of M_(111A)-I_(105A)) regulates theVG_(M119A) which increase the output impedance of the RGC-CM. This samediscussion is applicable to RGC_(102A), RGC_(104A), and RGC_(105A). As aresult, effectively, the output impedance and gain of CGAs and currentmirrors in FCTA is increased. Note that main amplifier'sR_(OUT)∝r_(o)×(r_(o)×g_(m))² and A_(V) is ∝R_(OUT)×g_(mi). Hence,A_(V)∝(r_(o)×g_(m))³, in the subthreshold region of operations for CMOS.To save current consumption, note that A_(V) can be increased by raisingonly I_(108A) and I_(111A), instead of increasing the currentconsumption of the whole amplifier, given that A_(V) is ∝R_(OUT)×g_(mi),and g_(mi) is the input stage transconductance that is about∝I_(108A)/V_(T)

Moreover, the DC common mode range of the main amplifier inputs(M_(120A) and M_(121A)) is expanded because, with the DC voltage shiftgenerated by the DCSCs in RGC_(102A) and RGC_(103A), the V_(108A) andV_(109A) can get closer to the rails. The DC voltage shift generated inDCSC of RGC_(103A) is approximately,VGS_(M113A)−VGS_(M115A)=ΔVGS≈VDS_(M113A)≈V_(T)×ln(r/s). Hence,V_(109A)≈VGS_(M111A)−V_(T)×ln(r) from the negative rail.

Additionally, the same type of FETs (e.g., NMOSFETs) are utilized in theCGAs function, the DC voltage shift function that expands the V_(OUT)span, and the auxiliary amplifier function that increases R_(OUT) of theRGC-CM. Utilizing the same type of FETs here, improves the consistencyof DC, AC, and transient performance over temperature, power supply, andprocess variations. FIG. 1H illustrates a worst case (WC) AC simulationof FIG. 1A that A_(V)˜105 dB, P_(M)˜45° is achievable. FIG. 1I uppergraph (i) is the WC transient simulation of FIG. 1A indicating thatSR˜2V/μS and t_(S)˜1 μS is achievable. Moreover, FIG. 1I lower graph(ii) indicates current consumption of ˜180 nano Ampere is achievable forFIG. 1A circuit.

In summary, besides increasing an amplifier's gain, the benefits ofutilizing plurality of RGC-CM_(100B) coupled with DCSC_(100B) in anamplifier are as follows. First, the same type (e.g., NMOSFET) FETs areused in each of the main amplifier's CGA, and RGC-CM's auxiliaryamplifiers plus DCSCs. Given that same type (e.g., NMOSFET) FET's deviceparameters match and track each other, therefore the FCTA's consistencyof DC and AC specifications and stability is improved over fabricationprocess, temperature, and power supply variations. Second, each of theamplifier's RGC_(101A) to RGC_(105A) are self biased (i.e., no need forseparate voltages to set its internal bias nodes) saving current and diespace. Third, the listed benefits of RGC-CM_(100B) coupled withDCSC_(100B) (see previous section) carry over to the amplifier,including small size, low current, and faster dynamic response. Fourth,the DC voltage shift provided in the RGC-CM_(100B) coupled withDCSC_(100B) widen the amplifier's input-output span closer to the rails.As stated before, for example, a 75 mV head-room expansion at the inputand output of each of the upper PMOS and the lower NMOS basedRGC-CM_(100B) coupled with DCSC_(100B), can expand the voltage span atthe input of the amplifier as well as at the high gain (impedance) nodeof FCTA by 150 mV or 15%, which is beneficial especially in the sub-1Vpower supply environment.

Section (V): Detailed Description of Amplifier (AMP_(100C)) in FIG. 1CUtilizing Plurality of RGC-CM_(100D) Coupled with ICMA_(100D), asIllustrated in FIG. 1D

FIG. 1C is a circuit schematic showing a embodiment of an folded cascodetransconductance amplifier (FCTA) that utilizes plurality ofRGC-CM_(100D) coupled with ICMA_(100D), illustrated in FIG. 1D. WhenRGC-CM_(100D) coupled with ICMA_(100D) is utilized in an amplifier, itcan expand its input and output voltage span and increase its outputimpedance, R_(OUT), which in turn increases the amplifier's gain, A_(V).Also, the benefits of RGC-CM_(100D) coupled with ICMA_(100D) are carriedover to improve the amplifier with low power and higher speed. Note thatalternative amplifier embodiments are possible such as an amplifier withNMOS input stage or complementary rail-to-rail input stages, otheramplifier topologies that are not FCTA, amongst others.

The connections of the elements in FIG. 1C are described as follows: Thebody terminal of all NMOSFETs in FIG. 1C are connected to node 2 whichis V_(SS), and the body terminals of PMOSFETs are connected to node 1which is V_(DD). The upper terminals of the bias current sourcesI_(101C), I_(103C), I_(105C), I_(107C), I_(109C), I_(111C), I_(111C),I_(M112C), and I_(M113C) are connected to node 1, which is V_(DD). Biascurrent sources I_(100C), I_(102C) I_(104C), I_(106C), and I_(108C) havetheir lower terminals connected to node 2, which is V_(SS). The sourceterminal of PMOSFETs M_(100C), M_(102C), M_(106C), M_(110C), M_(112C),M_(116C), and M_(120C) are connected to node 1 which is V_(DD). Thesource terminal of NMOSFETs M_(124C), M_(125C), M_(128C), M_(101C),M_(103C), M_(107C), M_(111C), M_(113C), M_(117C), and M_(121C) areconnected to node 2 which is V_(SS). Node 100C is the V_(IN+) terminalof the amplifier and connected to the gate terminal of M_(122C). Node101C is the V_(IN−) terminal of the amplifier and connected to the gateterminal of M_(123C). Node 102C is connected to the source terminals ofM_(122C), and M_(123C). Node 103C is connected to the drain terminal ofM_(122C), the drain terminal of M_(107C), the source terminal ofM_(105C), and the source terminal of M_(109C). Node 104C is connected tothe drain terminal of M_(123C), the drain terminal of M_(117C), thesource terminal of M_(115C), and the source terminal of M_(119C). Node105C is connected to the drain terminal of M_(100C), the gate terminalof M_(108C), and the upper terminal of I_(100C). Node 106C is connectedto the drain terminal of M_(102C), the gate terminal of M_(102C), thedrain terminal of M_(104C), and the upper terminal of I_(102C). Node107C is connected to the drain terminal of M_(108C), the drain terminalof M_(109C), the gate terminal of M_(106C), and the gate terminal ofM_(116C). Node 108C is connected to the drain terminal of M_(106C), andthe source terminal of M_(104C), the source terminal of M_(108C). Node109C is connected to the drain terminal of M_(110C), the gate terminalof M_(118C), and the upper terminal of I_(104C). Node 110C is connectedto the drain terminal of M_(112C), the gate terminal of M_(112C), thedrain terminal of M_(114C), and the upper terminal of I_(106C). Node111C is connected to the drain terminal of M_(116C), the source terminalof M_(114C), and the source terminal of M_(118C). Node 112C is connectedto the drain terminal of M_(120C), the gate terminal of M_(120C), thegate terminal of M_(114C), the gate terminal of M_(104C), and the upperterminal of I_(108C). Node 113C is connected to the drain terminal ofM_(124C), the gate terminal of M_(127C), and the lower terminal ofI_(111C). Node 114C is connected to the drain terminal of M_(125C), thegate terminal of M_(125C), the drain terminal of M_(126C), and the lowerterminal of I_(112C). Node 115C is connected to the drain terminal ofM_(127C), the lower terminal of I_(113C), the gate terminal of M_(128C),the gate terminal of M_(107C), and the gate terminal of M_(117C). Node116C is connected to the drain terminal of M_(128C), and the sourceterminal of M_(127C), the source terminal of M_(126C). Node 117C isconnected to the drain terminal of M_(101C), the gate terminal ofM_(109C), and the lower terminal of I_(101C). Node 118C is connected tothe drain terminal of M_(103C), the gate terminal of M_(103C), the drainterminal of M_(105C), and the lower terminal of I_(103C). Node 119C isconnected to the drain terminal of M_(111C), the gate terminal ofM_(119C), and the lower terminal of I_(105C). Node 120C is connected tothe drain terminal of M_(113C), the gate terminal of M_(113C), the drainterminal of M_(115C), and the lower terminal of I_(107C). Node 121C isthe high impedance high gain node or the output, V_(OUT), of theamplifier and connected to the drain terminal of M_(118C), and the drainterminal of M_(119C). Node 122C is connected to the drain terminal ofM_(121C), the gate terminal of M_(115C), the gate terminal of M_(105C),the gate terminal of M_(126C), the gate terminal of M_(121C), and thelower terminal of I_(109C).

Note that five RGC-CM_(100D) coupled with ICMA_(100D) are utilized inthe amplifier of FIG. 1C. The NMOS ones are RGC_(101C) made up ofM_(124C)-M_(125C)-M_(126C)-M_(127C)-M_(128C)-I_(111C)-I_(112C);RGC_(102C) made up ofM_(101C)-M_(103C)-M_(105C)-M_(107C)-M_(109C)-I_(101C)-I_(103C); andRGC_(103C) made up ofM_(111C)-M_(113C)-M_(115C)-M_(117C)-M_(119C)-I_(105C)-I_(107C). The PMOSones are RGC_(104C) made up ofM_(100C)-M_(102C)-M_(104C)-M_(106C)-M_(108C)-I_(100C)-I_(102C); andRGC_(105C) made up ofM_(110C)-M_(112C)-M_(114C)-M_(116C)-M_(118C)-I_(104C)-I_(106A). It wouldbe possible that the amplifier of FIG. 1A would function properlywithout RGC_(101C) and RGC_(104C), which can be substituted with diodeconnected cascodes. The RGC_(101C) and RGC_(104C) are added in theembodiment of FIG. 1C, in part, for better matching considerations andimproved AC performance. Note that, in order to avoid repeating how theamplifier is benefited by each of RGC_(101C) to RGC_(105C), theexplanation of one RGC-CM (e.g., RGC_(103C)) is deemed sufficient,regarding the function of RGC-CM_(100D) coupled with ICMA_(100D) in theFCTA.

For explanation regarding RGC-CM_(100D) coupled with ICMA_(100D) referto its detailed description. A general description is provided for morecontext in how utilizing RGC-CM_(100D) coupled with ICMA_(100D) canimprove the amplifier's performance. In the amplifier embodiment of FIG.1C, the V_(IN) is applied to a differential CSA, containing M_(122C) andM_(123C), whose current outputs feed a differential CGA, containingM_(109C) and M_(119C). Then, the differential current outputs of thisCGA feed a ‘current mirror’, made up of M_(106C) and M_(116C), to make asingle ended output, V_(OUT). The RGC-CM_(100D) coupled with ICMA_(100D)(of FIG. 1D) is chiefly utilized in the CGAs and ‘current mirrors’ ofthe FCTA of FIG. 1C, in order to increase the amplifier's outputimpedance, expand its input-output voltage span, which helps the FCTA'sdynamic response at low power.

Via M_(127C) and M_(128C), the current I_(113C)≈i is mirrored and scaledonto ID_(M107C)≈ID_(M117C)≈3i. The FCTA input currentsI_(M122C)≈I_(M123C)≈i (part of CSA) are fed into the source terminals ofM_(109C) and M_(119C), respectively, which are the differential inputsof the CGAs. The I_(103C)=2i and I_(107C)=2i flow through ICMAs ofRGC_(102C) and RGC_(103C), respectively, and passed onto M_(107C) andM_(117C), and in that order. Therefore, the operating currents of CGA'sare ID_(M109C)≈ID_(M107C)≈I_(103C)−ID_(M103C)−ID_(M122C)≈i andI_(M119C)≈ID_(M117C)−I_(107C)−ID_(M113C)−ID_(M123C)≈i. As explained inthe RGC-CM_(100D) coupled with ICMA_(100D) section, the auxiliaryamplifier function (performed by ICMA containing M_(115C), M_(113C),M_(111C), I_(107C), and I_(107C)) regulates the VG_(M119C) whichincrease the output impedance and gain of FCTA's CGA (containingM_(111C)), and thereby increases the R_(OUT) and gain (A_(V)) of theFCTA at node 121C. Note that main amplifier'sR_(OUT)∝r_(o)×(r_(o)×g_(m))² and A_(V) is ∝R_(OUT)×g_(mi). Hence,A_(V)∝(r_(o)×g_(m))³, with the amplifier FETs operating in thesubthreshold region. To save current consumption, note that amplifier'sgain can be increased by raising only I_(110C) and I_(113C), instead ofincreasing the current consumption of the whole amplifier, given thatA_(V) is ∝R_(OUT)×g_(mi), and g_(mi) is the input stage transconductancethat is roughly ∝I_(110C)/V_(T).

Moreover, the DC common mode range of the main amplifier inputs(M_(122C) and M_(123C)) is expanded. The M_(105C) and M_(115C) sourceterminals sense the 103C and 104C signals in current mode. The DC inputand output range of the amplifier is limited by V_(103C) and V_(104C),which can be predetermined by the gate voltages of M_(105C) andM_(115C), which are set by the scale and current of M_(121C). As such,VGS_(121C)−VGS_(115C)=VGS_(121C)−VGS_(105C)=ΔVGS≈VDS_(M117C)≈VDS_(M107C)≈V_(T)×ln(r/n),which enable the input and output voltages of the amplifier to can getmuch closer to the rails. Additionally, note that the same type of FETs(e.g., NMOSFETs) are utilized in the CGAs function, the DC voltage shiftfunction that expands the V_(IN) and V_(OUT) span, plus the auxiliaryamplifier function (performed by ICMA) that increases R_(OUT) ofRGC_(103C) and RGC_(105C) (and hence raises the A_(V) of the amplifier)which improves consistency of DC, AC, and transient performance overtemperature, power supply, and process variations

In summary, besides providing the extra gain, the benefits of utilizingplurality of RGC-CM_(100D) coupled with ICMA_(100D) in an amplifier areas follows. First, the same type (e.g., NMOSFET) FETs are used in eachof the main amplifier's common gate amplifiers (CGA), and RGC-CM'sauxiliary amplifiers (A_(AUX)) whose function is accomplished by theICMAs. Given that for example the NMOSFET device parameters match andtrack each other better, therefore the consistency and stability of themain amplifier's DC, AC, and transient specifications are improved overfabrication process, temperature, and power supply variations. If forexample CGA was based on NMOFETS, and A_(AUX) was based on mix ofPMOSFETs and NMSOFETs, then that would increase the risk ofinconsistencies in the amplifier's performance in the long run inmanufacturing. FIG. 1J illustrates a worst case (WC) AC simulation ofFIG. 1C that A_(V)˜110 dB (i.e., gain of over 315,000), and P_(M)˜40° isachievable with the amplifier utilizing the plurality of RGC-CM coupledwith ICMA. FIG. 1K upper graph (i) is the WC transient simulation ofFIG. 1C indicating that SR˜1.7 V/μS and t_(S)˜1.3 μS is achievable withthe amplifier of FIG. 1C utilizing the plurality of RGC-CM coupled withICMA. Moreover, FIG. 1K lower graph (ii) indicates current consumptionof ˜180 nano Ampere (i.e., one over one billion of an Ampere) isachievable with the amplifier of FIG. 1C utilizing the plurality ofRGC-CM coupled with ICMA. Second, the aforementioned benefits (see priorsection) of the same RGC-CM_(100D) coupled with ICMA_(100D) is utilizedin the amplifier in repeated (plurality of) instances, which carriesover its benefits to the amplifier including small size, low current,and higher speed. Third, the DC voltage shift provided in theRGC-CM_(100D) coupled with ICMA_(100D) widens the amplifier's rail torail span, which is beneficial especially in the sub-1V power supplyenvironment where every 10 mV, of extra voltage swing closer to therails, counts.

Section (VI): Detailed Description of Amplifier (AMP_(100E)) UtilizingPlurality of RGC-CM_(100F) Coupled with CSGA_(100F), as Illustrated inFIG. 1F

FIG. 1E is a circuit schematic showing a embodiment of a folded cascodetransconductance amplifier (FCTA) that utilizes plurality ofRGC-CM_(100F) coupled with CSGA_(100F) of FIG. 1F that is described inthe previous sections. When RGC-CM_(100F) coupled with CSGA_(100F) isutilized in a FCTA, it expands the input-output voltage span andincreases the amplifier's output impedance which in turn increases itsgain (A_(V)). Also, the other benefits of RGC-CM_(100F) coupled withCSGA_(100F), are carried over to improve the FCTA performance includingsmall size, low cost, low power, and higher speed.

Note that alternative amplifier embodiments may be possible such as anamplifier with NMOS input stage or complementary (PMOS and NMOS)rail-to-rail input stages, other amplifier topologies that are notfolded cascode transconductance, amongst others.

The connections of the elements in FIG. 1E are described as follows. Thebody terminal of all NMOSFETs in FIG. 1E are connected to node 2 whichis V_(SS), and the body terminals of PMOSFETs are connected to node 1which is V_(DD). Bias current sources I_(101E), I_(103E), I_(104E),I_(107E), I_(109E), I_(110E), I_(113E), I_(115E), and I_(119E) havetheir upper terminals connected to node 1, which is V_(DD). The lowerterminal of I_(100E), I_(102E), I_(106E), I_(108E), I_(117E), I_(105E),and I_(111E) are connected to node 2, which is V_(SS). Node 1 is alsoconnected to the drain terminal of M_(100E), the gate terminal ofM_(100E), the drain terminal of M_(110E), and the gate terminal ofM_(110E). Also, node 2 is connected to the drain terminal of M_(123E),the gate terminal of M_(123E), the drain terminal of M_(101E), the gateterminal of M_(101E), the drain terminal of M_(111E), and the gateterminal of M_(111E). The source terminal of M_(131E), M_(107E), andM_(117E) are connected to node 2 that is V_(SS). The source terminal ofM_(106E) and M_(116E) are connected to node 1 that is V_(DD). Node 101Eis connected to gate terminal of M_(120E), and it is the positive sideof the amplifier's input voltage terminal, or V_(IN)+. Node 102E isconnected to gate terminal of M_(121E), and it is the negative side ofthe amplifier's input voltage terminal, or V_(IN−). Node 103E isconnected to source terminal of M_(120E), the source terminal ofM_(121E), and the lower terminal of current source I_(112E). Node 104Eis connected to the drain terminal of M_(120E), the source terminal ofM_(109E), the drain terminal of M_(107E), and the gate terminal ofM_(105E). Node 105E is connected to the drain terminal of M_(121E), thesource terminal of M_(119E), the drain terminal of M_(117E), and thegate terminal of M_(115E). Node 106E is connected to source terminal ofM_(100E), the source terminal of M_(104E), the gate terminal ofM_(102E), and the upper terminal of current source I_(100E). Node 108Eis connected to drain terminal of M_(102E), the gate terminal ofM_(108E), and the upper terminal of I_(102E). Node 107E is connected tosource terminal of M_(102E), the drain terminal of M_(104E), and thelower terminal of I_(104E). Node 109E is connected to the sourceterminal of M_(108E), the drain terminal of M_(106E), and the gateterminal of M_(104E). Node 110E is connected to the drain terminal ofM_(109E), the drain terminal of M_(108E), the gate terminal of M_(106E),the gate terminal of M_(116E). Node 111E is connected to source terminalof M_(110E), the source terminal of M_(114E), the gate terminal ofM_(112E), and the upper terminal of current source I_(106E). Node 112Eis connected to the source terminal of M_(112E), the drain terminal ofM_(114E), and the lower terminal of I_(110E). Node 113E is connected todrain terminal of M_(112E), the gate terminal of M_(118E), the upperterminal of I_(108E). Node 114E is connected to the source terminal ofM_(118E), the drain terminal of M_(116E), and the gate terminal ofM_(114E). Node 115E is connected to the drain terminal of M_(119E), thedrain terminal of M_(118E), and it is the output of the amplifier,V_(OUT). Node 116E is connected to source terminal of M_(123E), thesource terminal of M_(127E), the gate terminal of M_(125E), and thelower terminal of current source I_(113E). Node 117E is connected todrain terminal of M_(125E), the gate terminal of M_(129E), and the lowerterminal of I_(115E). Node 118E is connected to source terminal ofM_(125E), the drain terminal of M_(127E), and the upper terminal ofI_(117E). Node 119E is connected to the drain terminal of M_(129E), thegate terminal of M_(131E), the gate terminal of M_(107E), and the gateterminal of M_(117E). Node 120E is connected to the source terminal ofM_(129E), the drain terminal of M_(131E), and the gate terminal ofM_(127E). Node 121E is connected to source terminal of M_(101E), thesource terminal of M_(105E), the gate terminal of M_(103E), and thelower terminal of current source I_(101E). Node 122E is connected todrain terminal of M_(103E), the gate terminal of M_(109E), and the lowerterminal of I_(103E). Node 123E is connected to source terminal ofM_(103E), the drain terminal of M_(105E), and the upper terminal ofI_(105E). Node 124E is connected to source terminal of M_(111E), thesource terminal of M_(115E), the gate terminal of M_(113E), and thelower terminal of current source I_(107E). Node 125E is connected todrain terminal of M_(113E), the gate terminal of M_(119E), and the lowerterminal of I_(109E). Node 126E is connected to source terminal ofM_(113E), the drain terminal of M_(115E), and the upper terminal ofI_(111E).

Note that there are five of RGC-CM_(100F) coupled with CSGA_(100F) thatare utilized in the amplifier embodiment of FIG. 1E. The lower ones onthe NMOS side of FCTA current mirrors are RGC_(101E) made up ofM_(123E)-M_(125E)-M_(127E)-M_(129E)-M_(131E)-I_(113E)-I_(115E)-I_(117E);RGC_(102E) made up of ofM_(101E)-M_(103E)-M_(105E)-M_(107E)-M_(109E)-I_(101E)-I_(103E)-I_(105E);and RGC_(103E) made up ofM_(111E)-M_(113E)-M_(115E)-M_(117E)-M_(119E)-I_(107E)-I_(109E)-I_(111E).On the PMOS side of FCTA current mirrors there are RGC_(104E) made up ofM_(100E)-M_(102E)-M_(104E)-M_(106E)-M_(108E)-I_(100E)-I_(102E)-I_(104E);and RGC_(105E) made up ofM_(110E)-M_(112E)-M_(114E)-M_(116E)-M_(118E)-I_(106E)-I_(108E)-I_(110E).It would be possible that the amplifier of FIG. 1E would functionproperly without RGC_(101E) and RGC_(104E), which can be substitutedwith diode connected cascodes. The RGC_(101E) and RGC_(104E) areutilized in the amplifier of FIG. 1E, in part, for systematic matchingconsiderations and improved AC performance. Note that, in order to avoidrepeating how the amplifier is benefited by each of RGC_(101E) toRGC_(105E), the explanation of one RGC-CM_(100F) coupled withCSGA_(100F) (e.g., RGC_(103E)) is deemed sufficient, regarding itsfunction in the FCTA.

For explanation regarding RGC-CM_(100F) coupled with CSGA_(100F), referto its detailed description. A general description is provided for morecontext in how utilizing RGC-CM_(100F) coupled with CSGA_(100F) canimprove the main amplifier's performance. The V_(IN) is applied to adifferential CSA, containing M_(120E) and M_(121E), whose currentoutputs feed a differential CGA, containing M_(109A) and M_(119A). Thedifferential current outputs of this CGA feed a current mirror functioncontaining M_(106E) and M_(116E), to make a single ended output,V_(OUT). For the FCTA during steady state, the current I_(119E)≈icontrols VGS_(M131E) that establishes ID_(M131E), which is mirrored andscaled onto ID_(M107E)≈ID_(M117E)≈2i. The FCTA's CSA input currentsI_(M120E)≈M_(121E)≈i are fed into the source terminals of M_(109E) andM_(119E), respectively, which are the differential inputs of the CGAs.At node 105E, the ID_(M120E)≈1i subtracts from ID_(M117E)≈2i whichprovides for ID_(M119E)≈1i. At node 104E, the ID_(M120E)≈i subtractsfrom ID_(M107E)≈2i which provides for ID_(M109E)≈1i. As explained in theRGC-CM coupled with composite amplifier in previous sections, theA_(AUX) (containing M_(113E)-I_(109E)) regulates the VG_(M119E) whichincreases R_(OUT) of RGC-CM in RGC_(103E). Similar discussion inapplicable to role of RGC_(102E), RGC_(104E), and RGC_(105E) in FCTAhere. As a result, effectively, the output impedance and gain of CGAsand current mirrors in FCTA are increased, and thereby the R_(OUT) andgain (A_(v)) of the FTCA are increased at node 115E. Note that mainamplifier's R_(OUT) is about ∝r_(o)×(r_(o)×g_(m))² and A_(V) is about∝R_(OUT)×g_(mi)∝(r_(o)×g_(m))³, in the subthreshold region ofoperations. To save current consumption, note that A_(V) can beincreased by raising only I_(112E) and I_(119E), instead of increasingthe current consumption of the FCTA, given that A_(v) is∝R_(OUT)×g_(mi), and g_(mi) is the input stage transconductance that isroughly ∝I_(112E)/V_(T). Moreover, the input DC common mode range of theFCTA (M_(120E) and M_(121E)) is expanded because, with the built-inoffsets (generated by the scaled M_(101E)-M_(105E) andM_(111E)-M_(115E)) in RGC_(102E) and RGC_(103E), the V_(104E) andV_(105E) can get closer to V_(SS). As noted earlier, this built-inoffsets in for example RGC_(103E) is approximately,VGS_(M111E)−VGS_(M115E)=ΔVGS≈VDS_(M117E)≈V_(T)×ln(r/s). Hence, by properscaling and operating current in M_(120E)-M_(121E), the inputs of FCTAcan span to the negative rail, V_(SS).

The voltage span of the output of FCTA is also improved. The built inoffset in PMOSFETs in RGC_(101E) to RGC_(103E) is chiefly a function ofV_(T) and mostly independent of PMOSFET device parameters, such asPMOSFET V_(TH). Similarly, the built in offset in NMOSFETs in RGC_(104E)and RGC_(105E) is chiefly a function of V_(T) and mostly independent ofNMOSFET device parameters, such as NMOSFET V_(TH). Therefore, thevoltages at nodes 104E, 105E, 109E, and 114E are mostly independent ofMOSFET device parameters, and track each other given that they aremostly a function of V_(T). Therefore, maximum V_(OUT) is approximately≤V_(DD)−V_(T)×ln(r/s)−VDS_(M118E-sat). Also, minimum V_(OUT) isapproximately ≤V_(SS)+V_(T)×ln(r/s)+VDS_(M119E-sat).

The amplifier is improved in consistency of DC, AC, and transientperformance over temperature, power supply, and process variations, inpart because the auxiliary amplifier (A_(AUX)) function in each ofRGC-CM_(100F) coupled with CSGA_(100F) is made of the same channel FETas the FCTA's CGA and current mirror. The fact that the built-in offsetfor all RGC-CM (coupled with composite amplifier) is mostly a functionof V_(T) also helps reduce systematic mismatches in the FCTA signal pathand helps improve FCTA performance over temperature, power supply, andprocess variations.

FIG. 1L illustrates a worst case (WC) AC simulation of FIG. 1E thatA_(V)˜110 dB, P_(M)˜30° is achievable with the FCTA after utilizing theplurality of RGC-CMs coupled with composite amplifiers. FIG. 1M uppergraph (i) is the WC transient simulation of FIG. 1E indicating thatSR˜3V/μS and t_(S)˜1 μS is achievable with the FCTA after utilizing theplurality of RGC-CMs coupled with composite amplifiers. Moreover, FIG.1M lower graph (ii) indicates current consumption of 180 nano Ampere isachievable with the FCTA after utilizing the plurality of RGC-CMscoupled with composite amplifiers.

In summary, besides providing the extra gain, the benefits of utilizingRGC with composite amplifier in the main amplifier include thefollowing. First, the same type of FETs are used in FCTA's CGA andcurrent mirrors and those used in RGC-CM's A_(AUX). The build-in offsetin RGC-CM is generated by PMOSFET and NMOSFET. However, the build-inoffset itself is roughly independent of either PMOSFET and NMOSFETdevice parameters such as CMOSFET V_(TH). The build-in offset is mostlya function of V_(T) which is highly predictable and insensitive toprocess variations. Therefore the FCTA's DC and AC specifications andstability is improved over fabrication process, temperature, and powersupply variations. Second, each of the amplifier's RGC_(101E) toRGC_(105E) are self biased (i.e., no need for separate voltages to setits internal bias nodes) saving current and die space. Third, theaforementioned benefits of the same RGC-CM_(100F) coupled withCSGA_(100F) that is utilized in the FCTA in repeated instances, carriesover to the FTCA the improvements in small size, low current, higherspeed, and wider output voltage span as well as consistency ofspecification performance over fabrication process, temperature, andpower supply variations. Fourth, the DC voltage shift provided in theRGC-CM_(100F) coupled with CSGA_(100F) expands the amplifier's DC commonmode input and output ranges closer to the rails. As noted earlier, forexample, a 75 mV head-room expansion at the input-output of each of theupper PMOS and the lower NMOS of RGC-CM_(100F) coupled with CSGA_(100F),can expand the voltage span at the input-output of the amplifier as wellas at the high gain (impedance) node of the FCTA by 150 mV or 15%, whichis beneficial especially in the sub-1V power supply environment.

Section (VII): Detailed Description of First Embodiment of an Amplifierof FIG. 2A Utilizing the First Floating Current Source (FCS_(200A))

FIG. 2A is a schematic circuit diagram of the embodiment illustrating anamplifier utilizing the first FCS (FCS_(200A)) shown in BLOCK:FCS_(200A) at the left bottom side of FIG. 2A. The embodiment disclosesFCS_(200A), which emulates the benefits of a FCS in the amplifiercurrent mirror network, enables the amplifier to operate at lowV_(DD)≤˜V_(GS)+2V_(DS), improves matching of upper and lower currentsindependent of the common mode voltage swings.

The amplifier of FIG. 2A is a conventional folded cascodetransconductance amplifier (FCTA) that utilizes FCS_(200A). As indicatedin prior sections, broadly speaking, a FCTA is composed of a CSA thatconverts the input differential voltage to differential currents whichare then fed onto a differential input and differential output CGA thatis fast. In a FCTA, the differential current outputs of CGA are thensteered onto a CM to generate a single output voltage gain at V_(OUT).The amplifier of FIG. 2A has two complementary inputs though, a PMOSFETinput and a NMOSFET input, operating in parallel in order to providerail-to-rail input operations which is consistent with the benefits ofthis disclosure. This arrangement is chosen so that either the PMOSFETinputs keep on operating when NMOSFETs run out of headroom near thenegative rail (V_(SS) or GND), or the NMOSFET inputs keep on operatingwhen PMOSFETs run out of headroom near the positive rail (V_(DD)). Onthe NMOSFET side of FCTA, the V_(IN) is applied to a differentialCSA_(N200A) containing M_(209A) and M_(211A). The differential currentoutputs of CSA_(N200A) then feed into a CGA_(P200A) containingM_(202A)-M_(214A) in conjunction with a current mirror (CM_(P200A))containing M_(200A)-M_(212A), the combination of which generates asingle ended output at node 212A. On the PMOSFET side of FCTA, theV_(IN) is applied to a differential CSA_(P200A) containing M_(208A) andM_(210A). The differential current outputs of CSA_(P200A) then feed intoa CGA_(N200A) containing M_(203A)-M_(215A) in conjunction with a currentmirror (CM_(N200A)) containing M_(201A)-M_(213A), combination of whichgenerates a single ended output at node 212A, which is the amplifier'shigh impedance, high gain output (V_(OUT)).

There are other amplifier configurations that can utilize this FCS. Onesuch example would be a FCTA with a g_(m) control circuit to keep theamplifier's input transconductance (g_(m)) constant across input voltagecommon mode range (VIN_(CM)). Another example would be FCTA thatutilizes regulated cascode current mirrors (RGCs) to improve theamplifier's performance, including increasing the gain of the amplifier.Note also that in FIG. 2A that the current source scale factors (e.g.,2×i=I_(205A) or 1×i=I_(201A), etc) and MOSFET scale factors (e.g., W/Lof M_(201A) versus W/L of M_(213A), or W/L of M_(204A) versus W/L ofM_(202A)) can be altered for different cost-performance goals, such asspeed versus power versus accuracy versus larger die which would costmore.

The connections of the elements in FIG. 2A are described as follows. Thebody terminal of all NMOSFETs in FIG. 2A are connected to node 2 whichis V_(SS), and the body terminals of PMOSFETs are connected to node 1which is V_(DD). The lower terminals of current sources I_(200A),I_(202A), and I_(205A) are connected to node 2. The upper terminals ofcurrent sources I_(201A), I_(203A), and I_(204A) are connected to node1. The source terminals of M_(200A), M_(206C), and M_(212A) areconnected to node 1. The source terminals of M_(201A), M_(207A), andM_(213A) are connected to node 2. Node 200A is connected to gateterminal of M_(206A), the drain terminal of M_(206A), the gate terminalof M_(204A), the gate terminal of M_(202A), and the gate terminal ofM_(214A). Node 201A is connected to the gate terminal of M_(207A), thedrain terminal of M_(207A), the gate terminal of M_(205A), the gateterminal of M_(203A), and the gate terminal of M_(215A). Node 202A isconnected to the gate terminal of M_(200A), the drain terminal ofM_(204A), the gate terminal of M_(212A), and the upper terminal ofcurrent source I_(200A). Node 203A is connected to the gate terminal ofM_(201A), the drain terminal of M_(205A), the gate terminal of M_(213A),and the lower terminal of current source I_(201A). Node 204A isconnected to the drain terminal of M_(209A), the drain terminal ofM_(200A), the source terminal of M_(204A), the source terminal ofM_(202A), and the drain terminal of M_(203A). Node 205A is connected tothe drain terminal of M_(208A), the drain terminal of M_(201A), thesource terminal of M_(205A), the source terminal of M_(203A), and thedrain terminal of M_(202A). Node 206A is connected to the sourceterminal of M_(208A), the source terminal of M_(210A), and the lowerterminal of current source I_(204A). Node 207A is connected to thesource terminal of M_(209A), the source terminal of M_(211A), and theupper terminal of current source I_(205A). Node 208A is connected to thedrain terminal of M_(211A), the drain terminal of M_(212A), and thesource terminal of M_(214A). Node 209A is connected to the drainterminal of M_(210A), the source terminal of M_(215A), and the drainterminal of M_(213A). Node 210A is connected to the gate terminal ofM_(208A), and the gate terminal of M_(209A), and it is the negativeinput terminal (V_(IN−)) of the amplifier. Node 211A is connected to thegate terminal of M_(210A), and the gate terminal of M_(211A), and it isthe positive input terminal (V_(IN)+) of the amplifier. Node 212A is thehigh impedance output (high gain node) of the amplifier, V_(OUT), whichis connected to the drain terminal of M_(215A), and the drain terminalof M_(214A). To be clear, the BLOCK FCS_(200A) contains the followingFETs: M_(200A), M_(202A), M_(204A), M_(206A), I_(200A), and I_(202A)plus M_(201A), M_(203A), M_(205A), M_(207A), I_(201A), and I_(203A).

One of the reasons for utilizing a FCS (or its equivalent function in anamplifier current mirror network) such as that of a FCTA, is to make thebiasing at the summing junction of the FCTA output more insensitive tothe common mode voltage swings in order to improve the amplifier'saccuracy. The prior art FCS circuit is illustrated in FIG. 2D,I_(M209D)+I_(M208D) flow into drains of M_(202D) and M_(203D), and thenonto M_(200D) and M_(201D), respectively, thus equalizing the operatingcurrents in M_(200D)and M_(201D) for the FCS. While this prior art FCScircuit in FIG. 2D is simple, but its minimum V_(DD) is restrictive. Theminimum V_(DD) of FIG. 2D is the greater ofVGS_(M204D)+VGS_(M206D)+VDS_(I202D) or˜VGS_(M205E)+VGS_(M207E)+VDS_(I203E), which is restrictive. The BLOCK:FCS_(200A) at the left bottom side of FIG. 2A, improves the FCSperformance including lowering its minimum power supply voltage, anddescribed as follows. For clarity, the operations of the FCS_(200A)independent of the amplifier is described first. Thus, ID_(M208A), andID_(M209A) are set to zero for now and non-idealities such as devicemismatches are set aside. The FCS_(200A) utilizes the first regulatingcircuit which is made up of M_(204A), and M_(206A), and current sourcesI_(200A), and I_(202A). Given that VGS_(M202A)=VGS_(M204A)⇒ID_(M202A)≈ID_(M204A)≈I_(N)=I_(200A). The second regulating circuitutilized in FCS_(200A) is made up of M_(205A), and M_(207A), and currentsources I_(201A) and I_(203A). Also, given thatVGS_(M205A)=VGS_(M203A)⇒ID_(M205A)≈ID_(M203A)≈I_(P)≈I_(201A). The KCL atnode 205A computes toID_(M205A)+ID_(M203A)+ID_(M202A)=ID_(M201A)≈2×I_(P)+ID_(M202A)≈2×I_(P)+I_(N).Similarly, the KCL at node 204A computes toID_(M204A)+ID_(M202A)+ID_(M203A)=ID_(M200A)≈2×I_(N)+ID_(M203A)≈2×I_(N)+I_(P).Setting I_(N)=I_(P)≈1×i, results in ID_(M200A)≈ID_(M201A)≈3×i, whichwill mirror onto M_(212A) and M_(213A), respectively. If there is amismatch of 10% and for example I_(P)≈0.9×i andI_(N)≈1×i⇒ID_(M200A)≈2.8×i and ID_(M201A)≈2.9×i⇒ID_(M200A)/ID_(M201A)≈97%. As such an error due to mismatch betweenI_(P) and I_(N) is roughly attenuated by a factor of 3 in thisFCS_(200A) embodiment. Now that the FCS_(200A) is explained, theoperations of the amplifier including the FCS_(200A) is described next.

The steady state operations of the FIG. 2A amplifier that utilizesFCS_(200A) is described first. With the amplifier operating under steadystate conditions, on the upper side of the amplifier,ID_(M209A)≈ID_(M211A)≈1×i. The first regulating circuit, containingM_(204A) and current source I_(M200A), regulate the gate voltage ofM_(200A), and the KCL operates on node 204A whereID_(M200A)≈ID_(M204A)+ID_(M202A)+ID_(M203A)+ID_(M209A)≈4×i. Given thatcurrents through M_(200A) and M_(212A) (≈2×i) are mirrored and scaled,the sum of currents at node 208A would result inID_(M214A)≈ID_(M212A)−ID_(M211A)≈i. Similarly, in steady stateconditions, the amplifier's inputs are balanced andID_(M208A)≈ID_(M210A)≈1×i. The second regulating circuit, containingM_(205A) and current source I_(M201A), regulate the gate voltage ofM_(201A), and the KCL operates on node 205A whereID_(M201A)≈ID_(M205A)+ID_(M203A)+ID_(M202A)+ID_(M208A)≈4×i. Given thatcurrents through M_(201A) and M_(213A) (≈2×i) are mirrored and scaled,the sum of currents at node 209A would result inID_(M215A)≈ID_(M213A)−ID_(M210A)≈i. Therefore, at the summing gain node212A of FCTA, operating current are in balance withID_(M214A)≈ID_(M215A)≈i.

With regards to node 204A, operating currents ID_(M202A)=ID_(M204A)≈iand ID_(M203A)≈i are held constant, setting aside non-idealities. As theKCL operates on node 204A, an input voltage change (Δv_(IN)) at the FCTAamplifier applied across M_(209A)-M_(211A) generates a current change(Δi_(N)) in ID_(M209A). This in turn would cause the first regulatingcircuit (containing M_(204A) and current source I_(M200A)) to regulatethe gate voltage of M_(200A) which results in the Δi_(N) to flow intoM_(200A), while ID_(M200A) is ‘current mirrored’ with ID_(M212A) (andscaled). Note that the dynamic response of this current mirror(M_(200A)-M_(212A)) is also improved. This is because the secondregulating circuit containing M_(204A) and current source I_(M200A) isconfigured as a common gate amplifier (CGA), which is very fast, andwhose output drives the gate terminals of M_(200A)-M_(212A).

Similarly, with regards to node 205A, as explained earlier, theoperating currents ID_(M203A)=ID_(M205A)≈i and ID_(M202A)≈i are heldconstant, setting aside non-idealities. As the KCL operates on node205A, an input voltage change (Δv_(IN)) at FCTA, applied acrossM_(208A)-M_(210A), generates a current change (Δi_(P)) in ID_(M208A).This in turn would cause the second regulating circuit (containingM_(205A) and current source I_(M201A)) to regulate the gate voltage ofM_(M201A) which would result in the Δi_(P) to flow into M_(M201A), whileID_(M201A) is ‘current mirrored’ with ID_(M213A) (and scaled). Note alsothat the dynamic response of this current mirror (M_(201A)-M_(213A)) isimproved. Similarly, this is because the second regulating circuitcontaining M_(205A) and current source I_(M201A) is configured as acommon gate amplifier (CGA), which is inherently fast, and whose outputdrives the gates terminals of M_(201A)-M_(213A).

The FCS_(200A) minimum V_(DD)≥V_(GS)+2V_(DS) is improved using fewertransistors and less current (versus prior art of FIG. 2D where minimumV_(DD)≥2V_(GS)+V_(IDS)). Note that, in utilizing FCS_(200A), the inputcommon mode span of the amplifier remains wide, given that node 203A,209A, and 204A, 208 a are set a V_(D)s above and below rails.

In summary, the FCS_(200A) block that is utilized in the amplifier ofFIG. 2A, operates as follows: the NMOSFET cascode current source(M_(201A) and M_(203A) cascoded on node 205A) and the PMOSFET cascodecurrent source (M_(200A) and M_(202A) cascoded on node 204A) arearranged such that the drain current of M_(202A) is fed into node 205Awhile the drain current of M_(203A) is fed into node 204A. Concurrently,a first regulating circuit regulates the gate terminal M_(200A), and asecond regulating circuit regulates the gate terminal M_(201A) such thatthe operating currents of M_(200A) and M_(201A) are substantiallyequalized, while the FCS_(200A) operates with low power supply voltages.

FIG. 2E upper graph (i) is the WC transient simulation of FIG. 2Aindicating that SR˜1V/μS and t_(S)˜1.5 μS is achievable with the FCTAutilizing the first FCS. Moreover, FIG. 2D middle graph (ii) indicatescurrent consumption of μ60 nano Ampere is achievable, and FIG. 2D bottomgraph (iii) indicates that with FCTA in unity gain configurationsubjected to a 1 volt (V) change in common mode input voltage followedby 1V change in output voltage would cause a μ40 Pico ampere (pA) changein the operating current of FCS, which is about 30 nA.

In conclusion, some of the benefits of utilizing FCS_(200A) in anamplifier may include one or more of the following. First, the FCS canoperate at lower power supply, using fewer transistors with lesscurrent, which is beneficial for the amplifier that utilizes this FCS.Second, the FCS can provides matching between upper and low cascodedcurrent sources which improves amplifier's performance, includinglowering its offset and noise. Third, given the regulating circuit ofFCS_(200A) is based on common gate amplifier (CGA) configuration, thedynamic response of the FCS is improved which improves the dynamicresponse of the amplifier that utilizes it.

Section (VIII): Detailed Description of Second Embodiment of anAmplifier of FIG. 2B Utilizing the Second Floating Current Source(FCS_(200B))

FIG. 2B is a schematic circuit diagram of the embodiment illustrating anamplifier utilizing the second FCS, which is depicted in BLOCK:FCS_(200B) at the left bottom side of FIG. 2B. Similar to the first FCSdiscussed in the previous sections, the disclosed FCS_(200B), emulatingthe function of a floating current mirror in the amplifier's currentmirror network, here that enables the amplifier to operate at lowV_(DD)≥˜V_(GS)+2V_(DS), improves speed, and improve accuracy by makingthe current matching more independent of common mode voltage swings.Refer to the description provided regarding the FCTA of FIG. 2A in theprevious section, which is similar to the amplifier herein FIG. 2B thatutilizes FCS_(200B). The FCS_(200B) illustrated in BLOCK: FCS_(200B) (atthe left bottom side of FIG. 2B) which also utilizes two cascode currentsources, PMOSFETs (M_(200B) and M_(202B)) and NMSOFETs (M_(201B) andM_(203B)). The upper and lower FETs (M_(202B) and M_(203B)) in the twocascodes are criss crossed by feeding the middle PMOSFET (M_(202B))drain current into the middle NMOSFET (M_(203B)) source terminal andconversely feeding the middle NMOSFET (M_(203B)) drain current into themiddle PMOSFET (M_(202B)) source terminal. Concurrently, regulatingcircuits regulate the gate voltages of the upper and lower FETs(M_(200B) and M_(201B)) in the two cascodes such that their operatingcurrents (ID_(M200B) and ID_(M201B)) are substantially equalized.

As noted in the prior sections, it would be possible that there areother amplifier configurations besides FCTA that can utilize thisFCS_(200B). Moreover, note that in FIG. 2B the current source scalefactors (e.g., 2×i=I_(205B) or 1×i=I_(201B), etc) and MOSFET scalefactors (e.g., W/L of M_(201B) versus W/L of M_(213B), or W/L ofM_(204B) and W/L of M_(202B) versus W/L of M_(216B) and W/L of M_(220B))can be altered in aiming for different performance goals, such as speedversus power versus accuracy.

The connections of the elements in FIG. 2B are described as follows. Thebody terminal of all NMOSFETs in FIG. 2B are connected to node 2 whichis V_(SS), and the body terminals of PMOSFETs are connected to node 1which is V_(DD). The lower terminals of current sources I_(200B),I_(202B), I_(205B), I_(206B), and I_(208B) are connected to node 2. Theupper terminals of current sources I_(201B), I_(203B), I_(204B),I_(207B), and I_(209B) are connected to node 1. Source terminals ofM_(200B), M_(206B), M_(212B), and M_(218B) are connected to node 1.Source terminals of M_(201B), M_(207B), M_(215B), and M_(219B) areconnected to node 2. Node 200B is connected to the gate terminal ofM_(206B), the drain terminal of M_(206B), the gate terminal of M_(202B),the gate terminal of M_(214B), and the gate terminal of M_(220B). Node201B is connected to the gate terminal of M_(207B), the drain terminalof M_(207B), the gate terminal of M_(203B), the gate terminal ofM_(215B), and the gate terminal of M_(22B). Node 202B is connected tothe drain terminal of M_(218B), the source terminal of M_(220B), and thesource terminal of M_(216B). Node 203B is connected to the drainterminal of M_(219B), the source terminal of M_(221B), and the sourceterminal of M_(217B). Node 204B is connected to the drain terminal ofM₂₂B, the gate terminal of M_(218B), and the upper terminal of I_(208B).Node 205B is connected to the drain terminal of M_(221B), the gateterminal of M_(219B), and the lower terminal of I_(209B). Node 206B isconnected to the drain terminal of M_(216B), the gate terminal ofM_(216B), the gate terminal of M_(204B), and the upper terminal ofI_(206B). Node 207B is connected to the drain terminal of M_(217B), thegate terminal of M_(217B), the gate terminal of M_(205B), and the lowerterminal of I_(207B). Node 208B is connected to the gate terminal ofM_(200B), the drain terminal of M_(204B), the gate terminal of M_(212B),and the upper terminal of current source I_(200B). Node 209B isconnected to the gate terminal of M_(201B), the drain terminal ofM_(205B), the gate terminal of M_(213B), and the lower terminal ofcurrent source I_(201B). Node 210B is connected to the drain terminal ofM_(200B), the source terminal of M_(204B), the source terminal ofM_(202B), the drain terminal of M_(203B), and the drain terminal ofM_(209B). Node 211B is connected to the drain terminal of M_(201B), thesource terminal of M_(203B), the source terminal of M_(205B), the drainterminal of M_(202B), and the drain terminal of M_(208B). Node 212B isconnected to the source terminal of M_(208B), the source terminal ofM_(210B), and the lower terminal of current source I_(204B). Node 213Bis connected to the source terminal of M_(209B), the source terminal ofM_(211B), and the upper terminal of current source I_(205B). Node 214Bis connected to the drain terminal of M_(211B), the drain terminal ofM_(212B), and the source terminal of M_(214B). Node 215B is connected tothe drain terminal of M_(210B), the source terminal of M_(214B), and thedrain terminal of M_(21B). Node 216B is connected to the gate terminalof M_(208B), and the gate terminal of M_(209B), and it is the negativeinput terminal (V_(IN−)) of the amplifier. Node 217B is connected to thegate terminal of M_(210B), and the gate terminal of M_(211B), and it isthe positive input terminal (V_(IN)+) of the amplifier. Node 218B is thehigh impedance output (high gain node) of the amplifier, V_(OUT), whichis connected to the drain terminal of M_(214B), and the drain terminalof M_(21B). To be clear, the BLOCK FCS_(200B) containing the following:M_(200B), M_(202B), M_(204B), M_(206B), M_(216B), M_(218B), M_(220B),I_(200B), I_(202B), I_(206B), on the top side and I_(208B) plusM_(201B), M_(203B), M_(205B), M_(207B), M_(217B), M_(219B), M_(221B),I_(201B), I_(203B), I_(207B), and I_(209B) on the bottom side.

For clarity and consistency with the prior sections, the operations ofthe FCS_(200B) is described first, independent of that of the amplifier.Thus, ID_(M208A), and ID_(M209A) are assumed to be zero for the purposeof this segment's description, and also non-idealities such as devicemismatches are set aside.

The FCS_(200B) top side utilizes the first regulating circuit made up ofM_(204B), M_(206B), M_(216B), M_(218B), and M_(220B), and currentsources I_(200B), I_(202B), I_(206B), and I_(208B). In this embodiment,the I_(200B)≈I_(206B)≈I_(208B)≈i are set as equals constant currentsources that bias VGS_(M204B)≈VGS_(M220B)≈VGS_(M216B). Applying the KVLto the voltage loop containingVGS_(M202B)=VGS_(M220B)−VGS_(M216B)+VGS_(M204B)≈VGS_(M220B). Therefore,ID_(M202B)≈ID_(M220)≈1i. The first regulating circuit is a current inputamplifier, where VGS_(M206B)−VGS_(M220B)+VGS_(M216B) establish theVG_(M204B). As such, M_(204B) and I_(200B) function like a common gateamplifier (CGA_(P200B)), which is fast. The output of this CGA_(P200B)regulates VGS_(M200B) at the node 208B until the KCL at node 210B issatisfied, which is whenID_(M200B)≈ID_(M202B)+ID_(M203B)+ID_(M204B)+ID_(M209B).

The second regulating circuit utilized in the bottom side of FCS_(200B)is made up of M_(205B), M_(207B), M_(217B), M_(219B), and M_(221B), andcurrent sources I_(201B), I_(203B), I_(207B), and I_(209B). Similar tothe top side, the I_(201B)≈I_(207B)≈I_(209B)≈1i are set as equalconstant current sources that bias VGS_(M205B)≈VGS_(M217B)≈VGS_(M221).Applying the KVL to the voltage loop containingVGS_(M203B)=VGS_(M221B)−VGS_(M217B)+VGS_(M205B)≈VGS_(M221B). Therefore,ID_(M203B)≈ID_(M221)≈1i. The second regulating circuit is a currentinput amplifier, where VGS_(M207B)−VGS_(M221B)+VGS_(M217B) establish theVG_(M205B). As such, M_(205B) and I_(201B) function like a common gateamplifier (CGA_(N200B)), which is fast. The output of this CGA_(N200B)regulates VGS_(M201B) at the node 209B until the KCL at node 211B issatisfied, which is whenID_(M201B)≈ID_(M203B)+ID_(M202B)+ID_(M205B)+ID_(M208B).

The steady state operations of the FIG. 2B amplifier that utilizesFCS_(200B) is described as follows. In steady state conditions, theamplifier's inputs are balanced and ID_(M208B)≈ID_(M210B)≈1×i. The firstregulating circuit, containing M_(204B) and I_(M200B), regulates thegate voltage of M_(200B) until KCL is satisfied by operating on node210B where ID_(M200B)≈ID_(M204B)+ID_(M203B)+ID_(M202B)+ID_(M209B) 4×i.Given that currents through M_(200B) and M_(212B) (≈2×i) are mirroredand scaled, the sum of currents at node 214B would result inID_(M214B)≈ID_(M212B)−ID_(M211B)≈i. Similarly, with the amplifieroperating under steady state conditions, on the upper complementary sideof the amplifier, ID_(M209A)≈ID_(M211A)≈1×i. The second regulatingcircuit, containing M_(205B) and I_(M201B), regulate the gate voltage ofM_(201B) until KCL is satisfied by operating on node 211B whereID_(M201B)≈ID_(M205B)+ID_(M203B)+ID_(M202B)+ID_(M208B)≈4×i. Given thatcurrents through M_(201B) and M_(213B) (≈2×i) are mirrored and scaled,the sum of currents at node 215B would result inID_(M215B)≈ID_(M213B)−ID_(M210B)≈i.

With regards to node 210B, operating currents ID_(M202B)=ID_(M204B)≈iand ID_(M203B)≈i are held constant, setting aside non-idealities. As theKCL operates on node 210B, an input voltage change (Δv_(IN)) acrossM_(209B)-M_(211B) generates a current change (Δi_(N)) in ID_(M209B) thatwould cause the first regulating circuit (containing M_(204B) andcurrent source I_(M200B)) to regulate the gate voltage of M_(200B). As aresult, Δi_(N) would flow into M_(200B), while ID_(M200B) is ‘currentmirrored’ with ID_(M212B) (and scaled). Note that the dynamic responseof this current mirror (M_(200B)-M_(212B)) is also improved. This isbecause this first regulating circuit containing M_(204B) and currentsource I_(M200B) is configured as a common gate amplifier (CGA), whichis inherently fast, and whose output drives the gate terminals of thecurrent mirror containing M_(200B)-M_(212B).

With regards to node 211B, the operating currentsID_(M203A)=ID_(M205A)≈i and ID_(M202A)≈i are held constant, settingaside non-idealities. While the KCL operates on node 211B, the inputvoltage change (Δv_(IN)) across M_(208B)-M_(210B) generates a currentchange (Δi_(P)) in ID_(M208B) which would cause the second regulatingcircuit (containing M_(205B) and current source I_(M201B)) to regulatethe gate voltage of M_(M201B). As a result, this Δi_(P) would flow intoM_(M201B), while ID_(M201B) is ‘current mirrored’ with ID_(M213B) (andscaled). Note also that the dynamic response of this current mirror(M_(201B)-M_(213B)) is improved. This is because this second regulatingcircuit containing M_(205B) and current source I_(M201B) is configuredas a common gate amplifier (CGA), which is very fast, and whose outputdrives the gates terminals of the current mirror containingM_(201B)-M_(213B). The FCS_(200B) minimum V_(DD)≥V_(GS)+2V_(DS) isimproved compared to the prior art of FIG. 2D where minimumV_(DD)≥2V_(GS)+V_(IDS). Note that, in utilizing FCS_(200B), the inputcommon mode span of the amplifier remains wide, given that nodes 203B,211B, 215B and 202B, 210B, 214B can be set as V_(DS sat) above and belowrails.

In summary, the FCS_(200B) block that is utilized in the amplifier ofFIG. 2B, operates as follows. The NMOSFET cascode current source(M_(201B) and M_(203B) cascoded on node 211B) and the PMOSFET cascodecurrent source (M_(200B) and M_(202B) cascoded on node 210B) arearranged such that the drain current of M_(202B) is fed into node 211Bwhile the drain current of M_(203B) is fed into node 210B. Concurrently,a first regulating circuit regulates the gate terminal M_(200B), and asecond regulating circuit regulates the gate terminal M_(201B) such thatthe operating currents of M_(200B) and M_(201B) are substantiallyequalized, while the FCS_(200B) operates with low power supply voltages.

FIG. 2F upper graph (i) is the WC transient simulation of FIG. 2Bindicating that SR˜1V/μS and t_(S)˜1 μS is achievable with the FCTAutilizing the second FCS. Moreover, FIG. 2E middle graph (ii) indicatescurrent consumption of ˜80 nano Ampere is achievable, and FIG. 2E bottomgraph (iii) indicates that with FCTA in unity gain configurationsubjected to a 1 volt (V) change in common mode input voltage followedby 1V change in output voltage would cause a˜5 Pico Ampere (pA) changein the operating current of FCS, which is about 40 nA.

In conclusion, some of the benefits of utilizing FCS_(200B) in anamplifier are the following. First, the FCS can operate at lower powersupply which frees the amplifier to operate with lower power supply aswell. Second, performance of FCS is improved, including its dynamicresponse, by separating nodes 207B from 201B and separating nodes 206Bfrom 200B, which helps roughly shield the CGAs used in FCS fromtransients on nodes 200B and 200B. Hence, the transient response of theamplifier, in which the FCS is utilized, can be improved. Third, the FCScan provides some matching between upper and low cascoded currentsources which improves amplifier's performance, including lowering itsoffset and noise of the amplifier in which the FCS is utilized. Fourth,given the regulating circuit of FCS_(200B) is based on common gateamplifier, the dynamic response of the FCS is improved which improvesthe dynamic response of the amplifier in which it is utilized.

Section (IX): Detailed Description of Third Embodiment of an Amplifierof FIG. 2C Utilizing the Third Floating Current Source (FCS_(200C))

FIG. 2C is a schematic circuit diagram of the embodiment illustrating anamplifier utilizing the third FCS, which is depicted in BLOCK:FCS_(200C) at the left bottom side of FIG. 2C. Similar to the FCSsdiscussed in previous sections, the disclosed FCS_(200B) emulates thefunction of floating current source in the amplifier's current mirrornetwork, here and enables the amplifier to operate at lowV_(DD)≥˜V_(GS)+2V_(DS), and improve accuracy by making current matchingmore independent of common mode swings.

As just noted, from a high level functional perspective, the embodimentof FCS_(200C) here is similar to that of FCS_(200A) and FCS_(200C)disclosed in the previous sections. The embodiment of the FCS_(200C) isillustrated in BLOCK: FCS_(200B) (at the left bottom side of FIG. 2C)which also utilizes two cascode current sources, PMOSFETs (M_(200C) andM_(202C)) and NMSOFETs (M_(201C) and M_(203C)). Here, the middle cascodeFETs (M_(202C) and M_(203C)) are arranged with their drain and sourceterminals crisscrossed. As such, the middle PMOSFET (M_(202C)) draincurrent is fed into the middle NMOSFET (M_(203C)) source terminal andconversely the middle NMOSFET (M_(203C)) drain current is fed into themiddle PMOSFET (M_(202C)) source terminal. Concurrently, regulatingcircuits regulate the gate voltages of the upper and lower FETs(M_(200C) and M_(201C)) in the two cascodes such that their operatingcurrents (ID_(M200C) and ID_(M201C)) are substantially equalized.

As noted in the prior sections, it is possible that there are otheramplifier configurations, besides a FCTA amplifier topology, that canutilize this FCS_(200C). Moreover, note that in FIG. 2C, it would bepossible that the current source scale factors (e.g., 2×i=I_(205C) or1×i=I_(209C), etc) and MOSFET scale factors (e.g., W/L of M_(201C)versus W/L of M_(213C), or W/L of M_(220C) versus W/L of M_(202C) or W/Lof M_(205C) and W/L of M_(217C)) can be altered in aiming for differentperformance goals, such as speed versus power versus accuracy.

The connections of the elements in FIG. 2C are described as follows. Thebody terminal of all NMOSFETs in FIG. 2C are connected to node 2 whichis V_(SS), and the body terminals of PMOSFETs are connected to node 1which is V_(DD). The lower terminals of current sources I_(200C),I_(202C), I_(205C), and I_(208C) are connected to node 2. The upperterminals of current sources I_(201C), I_(203C), I₂₀₄, and I_(209C) areconnected to node 1. The source terminals of M_(200C), M_(206C), M₂₁₂,M_(218C), M_(222C), and M_(224C) are connected to node 1. Sourceterminals of M_(201C), M_(207C), M_(213C), M_(219C), M_(223C), andM_(225C) are connected to node 2. Node 200C is connected to the gateterminal of M_(206C), the drain terminal of M_(206C), the gate terminalof M_(202C), the gate terminal of M_(214C), the gate terminal of M₂₂₀,and the upper terminal of I_(202C). Node 201C is connected to the gateterminal of M_(207C), the drain terminal of M_(207C), the gate terminalof M_(203C), the gate terminal of M_(215C), the gate terminal of M₂₂₁,and the lower terminal of I_(203C). Node 202C is connected to the gateterminal of M_(216C), the drain terminal of M_(218C), and the sourceterminal of M₂₂₀. Node 203C is connected to the gate terminal ofM_(217C), the drain terminal of M_(219C), and the source terminal ofM_(221C). Node 204C is connected to the gate terminal of M_(218C), thedrain terminal of M_(220C), and the upper terminal of I_(208C). Node205C is connected to the gate terminal of M_(219C), the drain terminalof M_(221C), and the upper terminal of I_(209 C). Node 206C is connectedto the gate terminal of M_(222C), the gate terminal of M_(224C), thedrain terminal of M_(224C), and the drain terminal of M_(204C). Node207C is connected to the gate terminal of M_(223C), the gate terminal ofM_(225C), the drain terminal of M₂₂₅, and the drain terminal ofM_(205C). Node 208C is connected to the gate terminal of M_(200C), thegate terminal of M_(212C), the drain terminal of M_(216C), and the drainterminal of M_(222C). Node 209C is connected to the gate terminal ofM_(201C), the gate terminal of M_(213C), the drain terminal of M_(217C),and the drain terminal of M_(223C). Node 210C is connected to the drainterminal of M_(200C), the source terminal of M_(202C), the drainterminal of M_(203C), the gate terminal of M_(204C), and the drainterminal of M_(209C). Node 211C is connected to the drain terminal ofM_(201C), the source terminal of M_(203C), the drain terminal ofM_(202C), the gate terminal of M_(205C), and the drain terminal ofM_(208C). Node 212C is connected to the source terminal of M_(204C), thesource terminal of M_(216C), and the upper terminal of I_(200 C). Node213C is connected to the source terminal of M_(205C), the sourceterminal of M_(217C), and the lower terminal of I_(201 C). Node 214C isconnected to the source terminal of M_(208C), the source terminal ofM_(210C), and the lower terminal of I_(204 C). Node 215C is connected tothe source terminal of M_(209C), the source terminal of M_(210C), andthe upper terminal of I_(205 C). Node 216C is connected to the drainterminal of M_(212C), the source terminal of M_(214C), and the drainterminal of M_(211C). Node 217C is connected to the drain terminal ofM_(213C), the source terminal of M_(215C), and the drain terminal ofM_(210C). Node 218C is connected to the gate terminal of M_(208C), thegate terminal of M_(209C), and it is the positive input voltage terminalof the amplifier V_(IN)+. Node 219C is connected to the gate terminal ofM_(210C), the gate terminal of M_(211C), and it is the negative inputvoltage terminal of the amplifier V_(IN−). Node 220C is connected to thedrain terminal of M_(214C), the drain terminal of M_(215C), and it isthe (high gain, high impedance) output voltage terminal of the amplifierV_(OUT). To be clear, the BLOCK FCS_(200C) is made up of the following:top side M_(200C), M_(202C), M_(204C), M_(206C), M_(216C), M_(218C),M_(220C), M_(222C), M_(224C), I_(200C), I_(202C), and the bottom sideI_(208C) plus M_(201C), M_(203C), M_(205C), M_(207C), M_(217C),M_(219C), M_(221C), M_(223C), M_(225C), I_(201C), I_(203C), and I_(209C)

For clarity and consistency with the prior sections, the operations ofthe FCS_(200C) is described first, independent of that of the amplifier.Thus, ID_(M208C), and ID_(M209C) are set to zero and non-idealities suchas device mismatches are set aside. The FCS_(200C) utilizes the firstregulating circuit, on the top side, made up of M_(206C), M_(218C), andM_(220C), and current sources, I_(202C), and I_(208C) plus the amplifierA_(N200C)(containing M_(204C), M_(266C), M_(226C), M_(222C), and currentsource I_(200C)). Here, A_(N200C)'s output regulates VGS_(M200C) untilits inputs are substantially equalized, which is whenVGS_(M202C)≈VGS_(M220C) that occurs when ID_(M202C)≈ID_(M220C).Therefore, ID_(M202C) ID_(M220)≈1i, and the KCL operating on node 210Cwould result in ID_(M200C)≈ID_(M202C)+ID_(M203C)+ID_(M209C).

On the complementary or bottom side, the FCS_(200C) utilizes the secondregulating circuit made up of M₂₀₇, M_(219C), and M_(221C), and currentsources, I_(203C), and I_(209C) plus the amplifier A_(P200C) (containingM_(205C), M_(217C), M_(225C), M_(223C), and current source I_(201C)).Here, A_(P200C)'s output regulates VGS_(M201C) until its inputs aresubstantially equalized, which is when VGS_(M203C)≈VGS_(M221C) thatoccurs when ID_(M203C)≈ID_(M221C). Therefore, ID_(M203C)≈ID_(M221)≈1i,and the KCL operating on node 211C would result inID_(M201C)≈ID_(M203C)+ID_(M202C)+ID_(M208C).

The steady state operations of the FIG. 2C amplifier that utilizesFCS_(200C) is described as follows. In steady state conditions, theamplifier's inputs are balanced and ID_(M208C)≈ID_(M210C)≈1×i. As statedearlier, the first regulating circuit containing A_(N200C) regulates thegate voltage of M_(200C) until KCL is satisfied by operating on node210C where ID_(M200C)≈ID_(M203C)+ID_(M202C)+ID_(M209C)≈3×i. Given thatcurrents through M_(200C) and M_(212C) (≈2×i) are mirrored and scaled,the sum of currents at node 216C would result inID_(M214C)≈ID_(M212C)−ID_(M211C)≈i. Similarly, with the amplifieroperating under steady state conditions, on the top or complementaryside of the amplifier, ID_(M209A)≈ID_(M211A)≈1×i. Also as statedearlier, the second regulating circuit containing A_(P200C) regulatesthe gate voltage of M_(201C) until KCL is satisfied by operating on node211C where ID_(M201C)≈ID_(M203C)+ID_(M202C)+ID_(M208C)≈3×i. Given thatcurrents through M_(201C) and M_(213C) (≈2×i) are mirrored and scaled,the sum of currents at node 217C would result inID_(M215C)≈ID_(M213C)−ID_(M210C)≈i.

Again while the KCL operates on node 210C, an input voltage change(Δv_(IN)) across M_(209C)-M_(211C) generates a current change (Δi_(N))in ID_(M209C) that would cause the first regulating circuit, containingA_(N200C), to regulate the gate voltage of M_(200C). As a result, Δi_(N)would flow into M_(200C), while ID_(M200C) is ‘current mirrored’ withID_(M212C) (and scaled). Similarly, with the KCL operating on node 211C,the input voltage change (Δv_(IN)) across M_(208C)-M_(210C) generates acurrent change (Δi_(P)) in ID_(M208C) that would cause the secondregulating circuit containing A_(P200C), to regulate the gate voltage ofM_(M201C). As a result, this Δi_(P) would flow into M_(M201C), whileID_(M201C) is ‘current mirrored’ with ID_(M213C) (and scaled).

The FCS_(200C) minimum V_(DD)≥V_(GS)+2V_(DS) is improved compared to theprior art of FIG. 2D where minimum V_(DD)≥2V_(GS)+V_(IDS). Note that, inutilizing FCS_(200C), the input common mode span of the amplifierremains wide, given that nodes 203C, 211C, 217C and 202C, 210C, 216C canbe set as V_(DS) above and below rails.

FIG. 2G upper graph (i) is the WC transient simulation of FIG. 2Cindicating that SR˜1.5V/μS and t_(S)˜1 μS is achievable with the FCTAutilizing the third FCS. Moreover, FIG. 2F middle graph (ii) indicatescurrent consumption of ˜60 nano Ampere is achievable, and FIG. 2F bottomgraph (iii) indicates that with FCTA in unity gain configurationsubjected to a 1 volt change in common mode input voltage followed by 1Vchange in output voltage would cause a˜70 Pico Ampere (pA) change in theoperating current of FCS, which is about 20 nA.

In summary, the FCS_(200C) block that is utilized in the amplifier ofFIG. 2C, operates as follows. The NMOSFET cascode current source(M_(201C) and M_(203C) cascoded on node 211C) and the PMOSFET cascodecurrent source (M_(200C) and M_(202C) cascoded on node 210C) arearranged such that the drain current of M_(202C) is fed into node 211Cwhile the drain current of M_(203C) is fed into node 210C. Concurrently,a first regulating circuit regulates the gate terminal M_(200C), and asecond regulating circuit regulates the gate terminal M_(201C) such thatthe operating currents of M_(200C) and M_(201C) are substantiallyequalized, while the FCS_(200C) operates with low power supply voltages.Some of the benefits of utilizing FCS_(200C) in an amplifier are thefollowing. First, the FCS can operate at lower power supply which freesthe amplifier to operate with lower power supply as well. Second, theFCS can provides better matching between upper and low cascoded currentsources which improves amplifier's performance, including lowering itsoffset and noise.

Section (X): Detailed Description of Amplifier Illustrated in FIG. 3A,Utilizing a First Noise Reduction Plus Speed Boost Circuit

FIG. 3A is a circuit schematic showing an amplifier (BLOCK 301A)utilizing the noise reduction plus speed boost circuit (BLOCK 300A). Oneof the contribution of this disclosure is a method of lowering theoutput noise of an amplifier by narrow-banding it, and thenreinvigorating the slower dynamic range of narrow-banded amplifier, byintroducing a bias current boost (that feeds the amplifier operatingcurrent) which is dynamically enabled when the input of the amplifierbecome imbalanced after receiving a large differential input transientsignal. The dynamic current boost aims to speed the amplifier's dynamicresponse by boosting not just the amplifier's slew rate but also itssettling time.

To improve noise the amplifier is narrow banded. To maintain ultra lowpower consumption, the steady state quiescent current consumption of theamplifier is kept at ultra low levels. To re-invigorate the dynamicresponse of the narrow banded amplifier back-up, the intermittent(dynamic) current consumption of the amplifier is increased. One way tonarrow band the amplifier is to connect a capacitor (Ce) to the highimpedance (high gain) node of the amplifier. Given the low currentconsumption of the amplifier, the added capacitor to narrow band theamplifier, also makes the dynamic response of the amplifier slow. Thedisclosed method to reinvigorate the narrow banded (for lowering thenoise of) amplifier, provides a ‘fast boost-on’ current and ‘slowboost-off’ current helps optimize for faster dynamic response.

The embodiment of FIG. 3A utilizes a folded cascode transconductanceamplifier (FCTA). This teaching arranges utilization of the same deviceswith similar device parameters (e.g., PMOSFETs) as inputs, compensationcapacitors (e.g., FET capacitors, or normal capacitors), and biasresistors in both FCTA and BLOCK_(300A), which helps with smootherdynamic (transition) response going into and coming out of boost mode,and more consistent yield to performance specifications over process andoperating condition variations.

The scale factors for FETs and current sources (e.g., M_(314A),M_(316A), I_(300A), I_(301A), I_(302A), and I_(304A),M_(306A)-M_(3108A), M_(304A)-M_(310A)) can be altered depending onfactors such as speed, and power consumption goal, amongst others. Forexample, with 0.01≤b≤1000000, 0.01≤t≤1000000, for the embodiment of FIG.3A, the current sources, i=10 nA where b=10 and t=20. Also, in thissegment of description the terms: amplifier, FCTA, or BLOCK_(301A) areused interchangeably. The terms: noise reduction plus current boostcircuit, current boost circuit, speed boost circuit, or BLOCK_(300A) arealso used interchangeably. This disclosure utilizes the current boostmethod with an amplifier configured as FCTA. However, there arealternative amplifier configurations that can utilize this disclosure'snoise reduction plus speed boost method that would be possible. Someexamples would be to apply the method of lowering noise plus boostingdynamic response of an amplifier by modifying BLOCK_(300A) to match inaccordance with an amplifier with NMOS input stage, or a FCTA withcascoded or regulated cascoded current mirrors, or complementary (PMOSand NMOS) rail-to-rail input stages, or amplifier topologies that arenot folded cascode transconductance, amongst others.

The connections of the elements in FIG. 3A are described as follows. Thebody terminal of all NMOSFETs in FIG. 3A are connected to node 2 that isthe V_(SS), and the body terminals of PMOSFETs are connected to node 1that is the V_(DD). The upper terminals of the bias current sourcesI_(300A), I_(302A), and I_(301A), are connected to node 1 that isV_(DD). The lower terminal of the bias current source I_(304A) isconnected to node 2 that is V_(SS). The lower terminal of the voltagesource V_(301A) is connected to node 2 that is V_(SS). The ground node(GND) or node 0 is connected to the first terminals of effectivecapacitances Ce_(300A) and Ce_(301A). The source terminal of PMOSFETsM_(314A), M_(316A), M_(305A), M_(307A), M_(309A), and M_(311A) areconnected to node 1 which is the positive supply voltage, V_(DD). Thesource terminal of NMOSFETs M_(304A), M_(306A), M_(308A), M_(317A),M_(319A), M_(321A), and M_(323A) are connected to the negative supplyvoltage, V_(SS). Node 300A is connected to the gate terminal of M_(300A)and the gate terminal of M_(301A). Also, node 300A is the V_(IN)+terminal of the FCTA and BLOCK_(300A). Node 301A is connected to thegate terminal of M_(302A) and the gate terminal of M_(303A). Also, node301A is the V_(IN−) terminal of the FCTA and BLOCK_(300A). Node 302 isconnected to the source terminal of M_(300A), the source terminal ofM_(302A), and the lower terminal of I_(300A). Node 303A is connected tothe source terminal of M_(301A), the source terminal of M_(303A), andthe drain terminal of M_(305A). Node 304A is connected to the drainterminal of M_(300A), the gate terminal of M_(304A), the drain terminalof M_(304A), and the gate terminal of M_(310A). Node 305A is connectedto the drain terminal of M_(301A), the drain terminal of M_(319A), andthe source terminal of M_(313A). Node 306A is connected to the drainterminal of M_(302A), the gate terminal of M_(306A), the drain terminalof M_(306A), and the gate terminal of M_(308A). Node 307A is connectedto the drain terminal of M_(303A), the drain terminal of M_(321A), andthe source terminal of M_(315A). Node 308A is connected to the drainterminal of M_(308A), and the source terminal of M_(310A). Node 309A isconnected to the drain terminal of M_(317A), the gate terminal ofM_(307A), the drain terminal of M_(307A), and the gate terminal ofM_(305A). Node 310A is connected to the drain terminal of M_(310A), thegate terminal of M_(312A), and the lower terminal of current sourceI_(302A). Node 311A is connected to the drain terminal of M_(313A), thegate terminal of M_(309A), the drain terminal of M_(309A), and the gateterminal of M_(311A). Node 312A is connected to the source terminal ofM_(312A), and the upper terminal of current source I_(304A). Node 313Ais the output, V_(OUT), of the FCTA and is connected to the drainterminal of M_(311A), the drain terminal of M_(315A), and the secondterminal of effective capacitance Ce_(301A). Node 314A is connected tothe drain terminal of M_(312A), the gate terminal of M_(314A), the drainterminal of M_(314A), the gate terminal of M_(316A), and the secondterminal of effective capacitance Ce_(300A). Node 315A is connected tothe gate terminal of M_(315A), the gate terminal of M_(313A), and thepositive terminal of voltage source V_(301A). Node 316A is connected tothe drain terminal of M_(316A), the drain terminal of M_(323A), the gateterminal of M_(323A), the gate terminal of M_(321A), the gate terminalof M_(319A), the gate terminal of M_(317A), and the lower terminal ofcurrent source I_(301A).

The BLOCK_(300A) contains M_(300A), M_(302A) (configured in CSA),M_(304A), M_(306A), M_(308A), M_(310A) (configured in minimum currentselector, MCS_(300A)), I_(302A), M_(312A), I_(304A), M_(314A), M_(316A),Ce_(300A), and Ce_(301A) (configured in providing the ‘boost on’ and‘boost off’ signal as well as shaping the ‘fast boost-on’ {e.g., slew}current and ‘slow boost-off’ {e.g., slow decay with one-time constant}current that is fed into the FCTA. The FCTA contains M_(301A), M_(303A)(input stage configured in CSA similar to that of the BLOCK_(300A)),M_(313A), M_(315A), (configured in CGA) M_(309A), M_(311A) (configuredin current mirror), M_(305A), M_(307A), M_(317A), M_(319A), M_(321A),M_(323A), V_(301A), and I_(301A) (configured in the operating currentand bias circuitry network for the FCTA).

Describing the details of the circuit in FIG. 3A is as follows to give ageneral description for more context in how utilizing the noisereduction plus speed boost circuit can improve the amplifier'sperformance. The V_(IN) is applied to a differential CSA, containingM_(319A) and M_(321A), has its current outputs feed the next CGA,containing M_(313A) and M_(315A). The differential outputs of this CGAfeed the CM, containing M_(309A) and M_(311A), to make a single-endedoutput, V_(OUT). The bandwidth of the amplifier is approximately

$ {\propto {{gm}_{M\; 301A}/{Ce}_{301\; A}} \propto {( \frac{i}{V_{T}} )/{Ce}_{M\; 301\; A}}} ).$All else equal, increasing the operating current ‘i’, speeds it up andvice versa, and increasing the effective capacitance at the highimpedance node of the FCTA at node 313A narrows bands and makes itsdynamic response slower and vice versa.

An amplifier's noise generally increases when the amplifier operateswith low currents. This disclosure reduces the output noise of theamplifier by narrow banding the amplifier at its high impedance node313A by increasing Ce_(301A), while keeping the steady state currentconsumption at ultra low levels. Because narrow banding the amplifier,slows its dynamic response (speed), then the operating current ‘i’ isboosted dynamically, which is triggered when the amplifier input(V_(IN)) stop tracking each other (and go off balance) after (V_(IN))receive a large transient signal. In this teaching, the dynamic responseof the amplifier, generally speaking, goes through two phases: the‘slewing time’ or ‘slew rate’ phase and the ‘settling time’ phase. Theslew rate (SR∝2i/Ce_(301A)) of FCTA is largely determined by theeffective capacitance (Ce_(301A)) at the amplifier's high impedanceoutput node 313, and the operating current of the amplifier, ‘i’, thatis scaled as in ID_(M315A) and ID_(M311A). The settling time of theFCTA, that is distinguished from SR for the purpose of this disclosure,is largely dominated by input stage gm_(M301A), Ce_(301A), and theamplifier's output impedance that is largely a function ofr_(ds)∝V_(A)/i. Because both the g_(m), ‘i’, and Ce of the noisereduction plus speed boost circuit, BLOCK_(300A), and that of theamplifier, FCTA, are arranged to be a function of similar deviceparameters on silicon, therefore the dynamic response of FCTA and BLOCK300A track each other more consistently over process and operatingcondition variations.

Although the gm_(M301A) of FCTA and gm_(M300A) of FCTA and BLOCK_(300A)at their input stages are similar, but an aspect of this disclosure isthat FCTA and BLOCK_(300A) are arranged to respond differently to theirinput voltage entering and exiting balance (i.e., in and out of steadystate conditions). This aspect of the disclosure, that will be describedshortly, helps a smoother and more consistent dynamic response in andout of the speed boost phase, over process and operating conditionvariations.

First, the steady state phase for FIG. 3A is described. When theamplifier and BLOCK_(300A) inputs are in steady-state, the boost signalremains off. The input stage of BLOCK_(300A) and that of the FCTA aresimilar, made up of the PMOSFET primary pair (M_(300A)-M_(302A) andM_(301A)-M_(303A), respectively). The input stage differential paircurrents ID_(M300A)≈I_(M302A)≈i are fed onto the minimum currentselector, MCS_(300A).

First, at a high level, lets describe the operations of the minimumcurrent selector (MCS_(300A)) composed of M_(304A) mirrored and scaledwith M_(310A) coupled with M_(306A) mirrored and scaled with M_(308A).When ID_(M304A)≈0→VGS_(M304A)≈0→VG_(M310A)≈0, and thus the outputcurrent of MCS_(300A) which is outputted via ID_(M310A)≈0, thus themin(ID_(M304A), ID_(M306A))≈ID_(M304A)≈0 is selected. Conversely, whenID_(M306A)≈0→VGS_(M306A)≈0→VG_(M308A)≈0, and thus the output current ofMCS_(300A) which is outputted via ID_(M310A)≈0, thus the min(ID_(M304A),ID_(M306A))≈ID_(M306A)≈0 is selected. In the steady state phase there iscurrent equilibrium, whenID_(M304A)≈i→VGS_(M304A)≈VGS_(i)→VG_(M310A)≈VGS_(i). Similarly inequilibrium, when ID_(M306A)≈i→VGS_(M306A)≈VGS_(i)→VG_(M308A)≈VGS_(i).Given that M_(310A) is scaled at 2× the size of M_(304A) and M_(308A) isscaled at 2× the size of M_(306A), and that M_(310A) is in series withM_(310A), and that VG_(M308A)≈VG_(M310A)≈VGS_(i), therefore the outputcurrent of MCS_(300A) which is outputted via ID_(M310A)≈i. Thus, in thesteady state phase when ID_(M304A)≈ID_(M306A)≈i, the FETs in MCS_(300A),including M_(304A), M_(306A), M_(308A), and M_(310A) are in balance andthe MCS_(300A) operates accordingly and min(ID_(M304A),ID_(M306A))≈ID_(M304A)≈ID_(M306A)≈i.

Therefore, in steady state given that the output of MCS_(300A) generatesi≈ID_(310A)>I_(302A)≈0.5i, then V_(310A) that controls the boost on-offsignal, pulls down (V_(310A)≈V_(SS)) and there is the boost-off phase.As a result, VG_(M312A)≈0→ID_(M312A)≈0→ID_(M314A)≈ID_(M316A)≈0. Thus, insteady state phase, the amplifier's bias current is supplied only byI_(301A), and with ID_(M316A)≈0 amplifier remains in the ‘boost off’phase.

In this segment, the ‘boost on’ phase is described. When the boost isinactive, ID_(M323A)≈I_(301A) establishes the ultra low quiescentoperating current of the amplifier. While the boost is active,BLOCK_(300A) dynamically generates a boost current that is added toID_(M323A) to boost the FCAT dynamic response. Here is how this happens.When large differential signal transients are applied at V_(IN), thenall or majority of the I_(300A) current 2i would flow through, forexample, M_(302A). Thus, ID_(M302A)≈ID_(M306A)≈2i andID_(M300A)≈ID_(M304A)≈0<<2i. Therefore, as noted earlier, thenMCS_(300A) operates accordingly and min(ID_(M304),ID_(M306))≈ID_(M304A)≈0<<2i. In this phase, MCS_(300A) output (flowingthrough M_(310A)) conduct near or at zero current, and hence V_(310A) ispulled up by I_(302A) (when V₃₁₀≈V_(DD)) and there is the boost-onphase. In the boost-on phase, all of I_(304A)≈b×i passes throughM_(312A) (that functions like a current switch) onto M_(314A) that ismirrored and scaled with M_(316A). Hence, ID_(M316A)≈b×t×i, which is theboost-on dynamic current that is added onto the amplifier bias networkvia M_(316A) to M_(323A). Note that the operating currents in the inputstage of FCTA (M_(301A)-M_(303A)) is boosted via the amplifier's biasnetwork that contains M_(323A)-M_(317A)-M_(307A)-M_(305A). As notedearlier, boosting the operating current in the amplifier's input stage,increases its ‘i’ and g₁, and thus speeds up the amplifier's dynamicresponse. Similarly, the operating currents in the high-impedancehigh-gain stage of FCTA (M_(313A)-M_(315A)-M_(319A)-M_(11A)) is boostedvia the amplifier's bias network that containsM_(323A)-M_(319A)-M_(321A). As noted earlier, boosting the operatingcurrent in the amplifier's high impedance stage, increases its ‘i’ andalso speeds up the amplifier's dynamic response, including its slewrate.

Before the dynamics of getting in and out of the current boost phase isdescribed, a clarification with respect to Ce_(301A) and Ce_(301A) is inorder. From a high level point of view, this disclosure is flexible inhow Ce_(301A) and Ce_(301A) are made. For example, Ce_(301A) andCe_(301A) can be made of similar devices such as standard capacitors,FET capacitors, or relying on C_(e) associated with the intrinsiccapacitances coupled to the 313A node of amplifier and the 314A node ofBLOCK_(300A). The dynamic response of the FCTA and BLOCK_(300A) aredesigned to match by using the same type of devices that impact theirdynamic response, including using similar C_(e) and running theamplifier and the noise reduction (plus speed boost) circuits withsimilar bias currents that track each other over process and operatingvariations. Matching the dynamic response of FCTA and BLOCK_(300A) helpswith smooth transient response in and out of boost plus consistency ofdynamic response across process and operating condition variations. TheCe_(301A) is the effective capacitance at the high gain high impedanceV_(OUT) node 313A of the amplifier that low pass filter filters thenoise. This Ce_(301A) can be made with FETs or regular capacitorsavailable in standard CMOS fabrication processes. Part of the Ce_(301A)can also be contributed by the intrinsic input capacitance associatedwith, for example, a buffer transistor drivers that could be coupled tothe main FCTA's V_(OUT) node 313A. Similarly, the Ce_(300A) is theeffective capacitance at node 314A, in BLOCK_(300A), that contributes toshaping the transient profile of the ‘fast boost-on’ current and ‘slowboost-off’ current that is injected into the amplifier. This Ce_(300A)capacitance can be made with FETs (e.g., PMOSFET capacitors), or regularcapacitors available in standard CMOS fabrication processes. Part or allof the effective capacitance, Ce_(300A), can also be contributed by theintrinsic input capacitance associated with, for example, Ce_(M314A) andCe_(M316A).

Next the dynamics of getting in and out of current boost phase isdescribed. When the boost-on signal is triggered, I_(304A)≈b×i slews thenode 314A. The ‘boost-on’ signal at node 310A turns on M_(312A), whichacts like a current switch, which causes I_(304A) to provide a ‘fastboost-on’ current onto M_(314A) that is mirrored and scaled up ontoM_(316A). Note that the speed at which node 314A slews (i.e., thedynamic profile of ‘fast boost-on’ current) is also a function ofCe_(314A), beside I_(304A).

For a better perspective about the dynamic response, note that while the‘fast boost-on’ current is still feeding the FCTA's current biasnetwork, the FCTA's SR remains boosted. At some point when theamplifier's inputs approach balance, BLOCK_(300A) triggers the boost offsignal (bringing down V_(310A) to V_(SS)) which shuts off the currentswitch M_(312A). Here, the flow of I_(304A) is closed off, and theamplifier enters the ‘boost-off’ phase when a decaying current continuesfeeding the FCTA bias network. When the boost off signal is triggeredand I_(304A) is cut off from M_(314A), the profile of ‘slow boost-off’current is a function of the Ze_(314A) and Ce_(300A), whereZe_(314A)∝1/gm_(M314A), which will be described further shortly.

An aspect of this disclosure is that the boost-off signals is generatedresponding to different levels of inputs equilibrium, although both FCTAand BLOCK_(300A) are receiving and mentoring the same input voltage. TheBLOCK_(300A) in this disclosure is arranged such that when its inputscan be coarsely equalized, it triggers the boost-off signal.BLOCK_(300A) can be arranged such that its inputs can be coarselyequalized in different ways. Some example are for BLOCK_(300A) to havelower gain than the main amplifier, or having the MSC_(300A) triggernode 310A when its currents (e.g., M_(304A) and M_(306A)) are coarselyequalized instead of accurately, or inserting a hysteresis in the inputsstage of BLOCK_(300A), or combination thereof, amongst other means.Another aspect of this disclosure is that after the boost off signal istriggered in BLOCK_(300A), subsequently there continues to be a slowdecay current (thus the term ‘slow boost-off’ current) that feeds theamplifier's operating bias current network, until the slow boost-offcurrent dies off. This arrangement improves the FCTA settling time. Theaforementioned attributes (boost off signal triggered when inputs areequalized coarsely, plus the slow-boost off current) improve theamplifier's settling time. As noted earlier, the ‘slow boost-off’current decay rate is roughly determined by node 314A's time constantset that is set approximately by the equivalent capacitance, Ce_(314A),and impedance, Ze_(314A), at node 314A. Note that in this embodimentCe_(314A) is dominated by effective capacitance Ce_(M314A) andCe_(M316A), and Ze_(314A) is dominated by approximately 1/gm_(M314A)computed in the saturation region, in light of the boosted transitioncurrents from subthreshold to saturation region during the intermittentboosted phase. As stated earlier, this embodiment enables the SR andsettling time of both the amplifier and BLOCK_(300A) (which provides andshapes the dynamic response profile of the ‘fast boost on’ and ‘slowboost-off’ currents for the FCTA) to approximately track and match eachother. As such, this trait benefits the amplifier with smooth dynamicsresponse (in and out of boost phases) and its consistency over processand operating condition variations.

Comparative simulations in FIG. 3C. depicts improvement in FIG. 3Aamplifier's output noise. Based on approximate device models, byutilizing the noise reduction and current boost circuit, the VO_(noise)of the amplifier at 10 Hz can be roughly 4 μV/√{square root over (Hz)}compared to roughly 8 mV/√{square root over (Hz)} for the comparableamplifier that does not utilize the noise reduction and current boostcircuit. Moreover, FIG. 300D depicts improvement in the FIG. 3Aamplifier's settling time (τ_(s)) with the amplifier utilizing the noisereduction and current boost circuit can be roughly 8 μs compared to μs110 for the comparable amplifier that does not utilize the noisereduction and current boost circuit.

In conclusion, here is a summary of some of the improvements to anamplifier's performance that can be attained by this disclosure. Firstthe amplifier's output noise is reduced by narrow banding the amplifier.Second, the static operating current consumption of the amplifier iskept ultra low. Third, the slowed dynamic response of the narrow-bandedamplifier (both slew rate and settling time) are reinvigorated andboosted. Fourth, lower noise, and most other attributes of the amplifierincluding its gain, bandwidth, static power consumption, common moderange, PSRR, and CMRR are generally not affected by the boost stage,since boost is only engaged dynamically when inputs experience largedifferential signals. Fifth, amplifier's input structure and that of theboost stage are substantially similar, and hence the boosting functionaccommodates the full common mode range and power supply span. Sixth, asnoted earlier, the AC, slew rate, and transient profiles of theamplifier and that of the boost circuit of BLOCK_(300A), shouldapproximately track each other over temperature and process variations.This is because generally, the amplifier and the boost stage's operatingcurrents, gain (e.g., ∝V_(A)/V_(T) ²), and input's 1/g_(m) (e.g.,∝V_(T)/i) roughly track each other, as do their poles in BLOCK_(300A)and FCTA are roughly a function of similar ∝1/g_(m), and ∝Ce_(FET) (orstandard capacitors), and ∝r_(o)≈V_(A)/i given that they are made ofsimilar device parameters for both BLOCK_(300A) and FCTA. Seventh, theswitching threshold of the amplifier's input stage from small signal tolarge signal needs to be large enough (e.g., offset mismatch between theboost and the amplifier input stages, ΔV_(OFS)) so that a false orpremature turning on of the boost function is avoided. This is the caseconsidering that in steady state (before boost-on phase), when theV_(IN) imbalance is first detected, the amplifier and boost stageinput's 1/gm_(PMOS) where i=10 s of nano-amperes. Note that theBLOCK_(300A) operating currents remain ultra-low, while the mainamplifier currents increase substantially in the boost-on phase to themicro-ampere range, which take the amplifier FETs out of thesubthreshold region. This is a transitory change in the gain as well asthe dynamic response of the amplifier while in the boost-on phase,compared to BLOCK_(300A) (that stays in subthreshold region) whilemonitoring and receiving the same input voltage as the amplifier. Notealso that, it is possible to provide some hysteresis at the input ofBLOCK_(300A) as guard-band against unwanted boost signal toggles.Eighth, to arrange for the boost-off signal to be triggered (setting inmotion the boost current decay) when the BLOCK_(300A) inputs arecoarsely equalized (as compared to the amplifier's inputs whose inputscontinue to converge towards finer balance and finer equalization), theamplifier's settling time is improved. Ninth, the maximum boost current,b×t×i, is fixed and is proportional to amplifier's static bias current,‘i’, which helps control peak dynamic current consumption. This traitalso facilitates the boost stage's peak speed to tracks that of the mainamplifier, over process, temperature, and operating variations. Notethat ‘i’ can be made independent of V_(TH) and mostly a function ofV_(T) and P_(PMOS) which are more tightly controlled in manufacturing,and helps with consistency of performance specification acrossmanufacturing variations. Tenth, care is taken to minimize dependence ofamplifier's specifications on multiple device parameters such asNMOSFET, V_(TH NMOS/PMOS), and N+/P+ resistors. Instead, amplifier'sspecifications mostly rely on PMOSFETs which dominate pertinent signalpaths, and this can help optimize yield and help lower noise (e.g.,PMOSFET 1/f noise<<NMOSFET 1/f noise) further.

Section XI: Detailed Description of Amplifier Illustrated in FIG. 3B,Utilizing a Second Noise Reduction Plus Speed Boost Circuit

FIG. 3B is a circuit schematic showing a amplifier utilizing the noisereduction plus speed boost circuit (BLOCK_(300B)). The embodimentillustrated in circuit schematic section in BLOCK_(300B) (of FIG. 3B)utilizes a Loser Take All LTA_(300B) circuit (compared to the MCS_(300A)that was utilized in BLOCK_(300A) of FIG. 3A). All else is substantiallysimilar and the description of the disclosure is interchangeablyapplicable between FIG. 3B and that of FIG. 3A. Beyond what is alreadydescribed in the previous section, the description here provides thedescription of elements, and detailed explanation of the LTA_(300B),with a summary in its conclusion.

Similar to the teaching in previous section, the disclosure here is amethod of lowering the output noise of an amplifier by band passing itwhile keeping its static current consumption ultra low, and concurrentlyspeeding up the amplifier's dynamic response, by boosting theamplifier's operating current in the face of the amplifier's inputsreceiving a large transient signal. The noise reduction and speed boostcircuit of BLOCK_(300B) utilizes a LTA_(300B) to detect when inputs areimbalanced. Similar to FIG. 3A, the circuit of FIG. 3B speeds up theamplifier's dynamic response by boosting both its SR as well as its'settling time. The scale factors for FETs and current sources (e.g.,M_(304B), M_(310B), M_(314B), M_(316B), I_(300B), I_(301B), I_(302B),and I_(304B)) can be altered depending on factors such as speed, powerconsumption goal, amongst others. For example, with 0.01≤b≤1000000,0.01≤t≤1000000, for an embodiment of FIG. 3B, the current sources, i=10nA where b=10, and t=20. This disclosure utilizes the current boostmethod with an amplifier configured as FCTA. However, there arealternative amplifier configurations that can utilize this disclosure'snoise reduction plus speed boost method that would be possible. Someexamples would be to apply the method of lowering noise plus boostingdynamic response of an amplifier by modifying BLOCK_(300B) to match inaccordance with an amplifier with NMOS input stage, or a FCTA withcascoded or regulated cascoded current mirrors, or complementary (PMOSand NMOS) rail-to-rail input stages, or amplifier topologies that arenot FCTA, amongst others. It would be also be possible to utilizevariations of MCS_(300A) or LTA_(300B) or combination thereof to detectan imbalance at the input of an amplifier, including for exampleutilizing Winner Takes All (WTA) function, amongst others.

The connections of the elements in FIG. 3B are described as follows. Thebody terminal of all NMOSFETs in FIG. 3B are connected to node 2 that isthe V_(SS), and the body terminals of PMOSFETs are connected to node 1that is the V_(DD). The upper terminals of the bias current sourcesI_(300B), I_(302B), and I_(301B), are connected to node 1 that isV_(DD). The lower terminal of the bias current source I_(304B) isconnected to node 2 that is V_(SS). The lower terminal of the voltagesource V_(301B) is connected to node 2 that is V_(SS). The ground node(GND) or node 0 is connected to the first terminals of effectivecapacitances Ce_(300B) and Ce_(301B). The source terminal of PMOSFETsM_(314B), M_(316B), M_(305B), M_(307B), M_(309B), and M_(311B) areconnected to node 1 which is the positive supply voltage, V_(DD). Thesource terminal of NMOSFETs M_(304B), M_(304B)′, M_(306B), M_(306B)′,M_(308B), M_(310B), M_(317B), M_(319B), M_(321B), and M_(323B) areconnected to the negative supply voltage, V_(SS), which is node 2. Node300B is connected to the gate terminal of M_(300B) and the gate terminalof M_(301B). Also, node 300B is the V_(IN)+ terminal of the FCTA andBLOCK_(300B). Node 301B is connected to the gate terminal of M_(302B),the gate terminal of M_(302B)′, and the gate terminal of M_(303B). Also,node 301B is the V_(IN−) terminal of the FCTA and BLOCK_(300B). Node 302is connected to the source terminal of M_(300B), the source terminal ofM_(302B), the source terminal of M_(302B)′, and the lower terminal ofI_(300B). Node 303B is connected to the source terminal of M_(301B), thesource terminal of M_(303B), and the drain terminal of M_(305B). Node304B is connected to the drain terminal of M_(300B), the gate terminalof M_(304B), the drain terminal of M_(304B), and the gate terminal ofM_(304B)′. Node 305B is connected to the drain terminal of M_(301B), thedrain terminal of M_(319B), and the source terminal of M_(313B). Node306B is connected to the drain terminal of M_(302B), the drain terminalof M_(304B)′, the gate terminal of M_(306B), the drain terminal ofM_(306B), and the gate terminal of M_(306B)′. Node 307B is connected tothe drain terminal of M_(303B), the drain terminal of M_(321B), and thesource terminal of M_(315B). Node 308B is connected to the drainterminal of M_(306B)′, the drain terminal of M_(308B), the gate terminalof M_(308B), and the gate terminal of M_(310B). Node 309B is connectedto the drain terminal of M_(317B), the gate terminal of M_(307B), thedrain terminal of M_(307B), and the gate terminal of M_(305B). Node 310Bis connected to the drain terminal of M_(310B), the gate terminal ofM_(312B), and the lower terminal of current source I_(302B). Node 311Bis connected to the drain terminal of M_(313B), the gate terminal ofM_(309B), the drain terminal of M_(309B), and the gate terminal ofM_(311B). Node 312B is connected to the source terminal of M_(312B), andthe upper terminal of current source I_(304B). Node 313B is the output,V_(OUT), of the FCTA and is connected to the drain terminal of M_(311B),the drain terminal of M_(315B), and the second terminal of effectivecapacitance Ce_(301B). Node 314B is connected to the drain terminal ofM_(312B), the gate terminal of M_(314B), the drain terminal of M_(314B),the gate terminal of M_(316B), and the second terminal of effectivecapacitance Ce_(300B). Node 315B is connected to the gate terminal ofM_(315B), the gate terminal of M_(313B), and the positive terminal ofvoltage source V_(301B). Node 316B is connected to the drain terminal ofM_(316B), the drain terminal of M_(323B), the gate terminal of M_(323B),the gate terminal of M_(321B), the gate terminal of M_(319B), the gateterminal of M_(317B), and the lower terminal of current source I_(301B).

The BLOCK_(300B) contains M_(300B), M_(302B), and M_(302B) (configuredin CSA), M_(304B), M_(304B)′, M_(306B), M_(306B)′, M_(308B), M_(310B)(configured in Looser Take All, LTA_(300B)), I_(302B), M_(312B),I_(304B), M_(314B), M_(316B), Ce_(300B), and Ce_(301B) (configured inproviding the ‘boost on’ and ‘boost off’ signal as well as shaping the‘fast boost-on’ (e.g., slew) current and ‘slow boost-off’ {e.g., slowdecay with one-time constant} current that is fed into the FCTA. TheFCTA contains M_(301B), M_(303B) (input stage configured in CSA similarto that of the BLOCK_(300B)), M_(313B), M_(315B), (configured in CGA)M_(309B), M_(311B) (configured in current mirror), M_(305B), M_(307B),M_(317B), M_(319B), M_(321B), M_(323B), V_(301B), and I_(301B)(configured in the operating current and bias network circuitry for theFCTA).

Here, description of the operations Loser Take All, LTA_(300B) (which issimilar in function to that of prior section regarding MCS_(300A)) isprovided, ignoring non-idealities such as mismatches. The boundaryconditions are described first. When large signals are applied to V_(IN)and input FETs M_(300B)-M_(302B)-M_(302B)′ become imbalanced. First, thecase is described when all of I_(300B)=4×i flows through M_(300B), andID_(M302B)≈ID_(M302)′≈0<<4×i. Therefore, ID M_(304B)≈2×ID_(M304B)′≈4×i.Given that ID_(M302B)≈0 andID_(M304B)′≈2×i→V_(306B)≈0→VGS_(M306B)≈0→ID_(M006B)≈ID_(306B)′≈0. Giventhat ID_(M302B)′≈0 and ID_(M306B) ′≈0→ID_(M308B)≈0≈ID_(M310B). Thus theoutput current of LTA_(300B) is outputted via ID_(M310B)≈0, thus theloser(ID_(M300B), ID_(M302B&B)′)≈ID_(M302B&B)′≈0 takes all. Note thatID_(M302B&B)′ denotes ID_(M302B), ID_(M302B)′ that are equal (having thesame VGS) whose sum is 2i during steady state.

Next, considering the case where the large signal V_(IN) causes animbalance such that all of I_(300B)=4×i flows through M_(302B) andM_(302B)′. Here, ID_(M300B)≈0<<4×i and ID_(M302B)≈ID_(M302B)′≈2×i.Therefore, ID_(M304B)≈2×ID_(M304B)≈0<<4×i. With ID_(M302B)≈2×i andID_(M304B)′≈0→ID_(M306B)≈2×i≈ID_(M306B)′. With ID_(M302B)′ 2×i andID_(M306B)′ ≈2×i, therefore no current is left for M_(308B). Thus,ID_(M308B)≈0≈ID_(M310B). Therefore the output current of LTA_(300B) isoutputted via ID_(M310B)≈0, thus the loser(ID_(M300B),ID_(M302B&B)′)≈ID_(M300B)≈0 takes all.

In summary, when there is an imbalance (in either direction) as a resultof a large signal V_(IN), the LTA_(300B) generates a zero currentthrough ID_(M310B)≈0. As such I_(302B)=1×i pulls up on node 310B(V_(310B)≈V_(DD))→M_(312B) turns on and passes I_(304B)=b×i ontoM_(314B) whose current is mirrored and scaled up onto M_(316B) withID_(M314B)=b×t×i. This ID_(M314B) is the boost-on current that is addedto the FCTA quiescent current, I_(304B), to feed the FCTA current biasnetwork through M_(323B).

Next considering the steady state, or static, conditions when theamplifier is regulated its input FETs M_(300B)-M_(302B)-M_(302B)′ are inbalance. In steady state, ID_(M300B)≈2×ID_(M302B)≈2×ID_(M302B)′≈2×i.Therefore, ID_(M304B)≈2×ID_(M304B)′ ≈2×i and ID_(M302B)≈i. Therefore,there is no current left for M_(306B)→ID_(M306B)≈ID_(M306B)′≈0. WithID_(306B)′≈0, then all of ID_(M302B′)≈i flows through M_(308B) which ismirrored and scaled up 2× into M_(310B) and thus2×ID_(M308B)≈ID_(M310B)≈2×i. Thus the output current of LTA_(300B) isoutputted via ID_(M310B)≈2×i since there are no loser, but equals per sehere, the loser(ID_(M300B), ID_(M302B&B)′)≈ID_(M300B)≈ID_(M302B&B) ′≈2×itakes all. Therefore, when the amplifier's inputs arrive at balance insteady state conditions, the output current of LTA_(300B) generates aID_(M310B)≈2×i (½ the tail current of the input stage). As such,ID_(M310B) pulls down on I_(302B)=1×i which takes node 310B down(V_(310B)≈V_(SS))→M_(312B) turns off and blocks I_(304B)=b×i fromflowing into M_(314B). Thus, ID_(M314B)≈0≈ID_(M316B). As a result, theultra low static FCTA quiescent current, I_(304B), feeds M_(323B) insteady state conditions with no boost current.

In summary, FIG. 3B is a circuit schematic based on a method of reducingan amplifier's output noise by narrow-banding the amplifier, which slowsits dynamic response, while independently reinvigorating the amplifier'snarrow-banded dynamic response via boosting both its' slew rate andsettling time. This is accomplished as follows. PMOSFET or standardcapacitors can be used at the high impedance node to low pass filter theamplifier's output noise. A loser take all (LTA) circuit is utilized todetect and trigger a boost on and boost off signal, when the amplifier'sinputs approximately go in and out of balance. A boost function isenabled, in response to large differential transient input signals, byinjecting a dynamic current into the amplifiers pre-existing static biascurrent network, which at first rapidly speeds up the amplifier's slewrate. When inputs approximately approach balance, the boost signal istriggered off. But here, a slow and decaying current continues feedingthe amplifier's bias current network, until the decaying current fadesoff. This decaying current (which is initially sizable enough whencompared to the otherwise statically ultra low current operant in theun-boosted mode) roughly follows a single pole trajectory, which helpsspeed up the amplifier's settling time.

As explained earlier, while reinvigorating the SR and settling time, theprice to pay for lowering the amplifier's noise, is the increaseddynamic current consumption (intermittent power consumption). After theamplifier's inputs receive a large signal transient, and the boost-onsignal is triggered, the ‘fast boost-on’ current kick starts speeding upthe amplifier's dynamic response by boosting up its slew rate, whichmoves the amplifier back on the path towards balancing its inputs aheadof boost-off signal trigger. To save on the intermittent powerconsumption, the boost off signal can be triggered sooner which cuts offthe slewing current boost sooner. However, a ‘slow boost-off’arrangement continues injecting a dynamic decaying current into theamplifier bias network (which eventually levels off to zero) to speed upthe amplifier's settling time. This is accomplished, in part, asexplained previously above by arranging BLOCK_(300B) to initiate theboost-off signal when its inputs are not equalized in the same size andfashion as that of the inputs of FCTA (e.g., the amplifier hasapproached but not fully regulated).

The whole amplifier operates in the subthreshold region when it isun-boosted, but its' input, transconductance gain, and (would be) outputbuffer stages would become faster by receiving the boosted dynamiccurrent and transition in and out of saturation during the intermittentboosted phases. It is also of note that PMOSFETs can be used at inputstages and also as active resistors to set the operating bias currentsfor the whole amplifier (including, input stages, gain stage, the boostcircuit, and output stage), which can establish both their input stage'stransconductance (gm_(PMOS)). Moreover, PMOSFET or normal capacitors canbe utilized to set the dominant poles of each of the amplifier gainstage and that of the boost stage. In other words, dynamic response ofboth the FCTA and BLOCK_(300B) can be largely proportional to the same(operating current) ‘i’ and (dominant effective capacitance) ‘C_(e)’,and same kind of device parameters. Hence, the dynamic response in andout of boost track each other, and follow a reasonably smooth and stablepassage, in and out of boost phases. Moreover, the profiles of theamplifier and the boost stage dynamic response can be more consistentfrom lot-to-lot in manufacturing, across operating variations.

Section (XII): Describing a Prior Art Amplifier Gain Stage Illustratedin FIG. 4A, Coupled with (Inverting) Buffer Driver

Generally a buffer amplifier, as depicted in FIG. 5, contains anamplifier section coupled with a buffer driver section. The amplifiersection amplifies the signals applied to its inputs, and has high inputimpedance and high output impedance and is sometimes referred to gainstage. Buffer driver section has high input impedance and low outputimpedance with its outputs capable of sinking and sourcing currents todrive external loads, such as resistors, capacitors, inductors or acombination thereof. Combination of amplifier and buffer driver makes abuffer amplifier that can amplify signals applied to its inputs. Abuffer driver also has high input impedance, low output impedance, andits output is capable of sinking and sourcing currents to drive externalloads. Therefore, there may be formed a buffer amplifier (AMP₄₀₀)comprised of a amplifier gain stage of FIG. 4A, coupled with a bufferdriver (e.g., inverting buffer driver). The (inverting) buffer driverscan be anyone of the disclosed BUF_(400B), BUF_(400C), and BUF_(400D)embodiments illustrated in FIG. 4B, FIG. 4C, or FIG. 4D (respectively),which will be discussed in the next sections. Note that AMP_(400B) isformed by amplifier gain stage of FIG. 4A coupled with BUF_(400B). TheAMP_(400C) is formed by amplifier gain stage of FIG. 4A coupled withBUF_(400C). The AMP_(400D) is formed by amplifier gain stage of FIG. 4Acoupled with BUF_(400D). The connection of the elements in FIG. 5 aredescribed as follows. Node 500A is the negative input terminal of theBuffer Amplifier that is connected to the positive input terminal of theAmplifier (that provides the gain stage). Node 501A is the positiveinput terminal of the Buffer Amplifier that is connected to the negativeinput terminal of the Amplifier (that provides the gain stage). Node502A is the output terminal of the Amplifier (that provides the gainstage) that is connected to the input terminal of the (inverting) bufferdriver (that provides current drive capability to an external load).Node 503A is the output of the Buffer Driver that is also the output ofthe Buffer Amplifier (that provides the Buffer Amplifier current drivecapability to an external load). Note that it would be obvious for onein the art to modify the disclosed teaching wherein the output of theAmplifier can be composed of 2 outputs (instead of one output at node502A) that would feed 2 inputs of a Buffer Driver (instead of one inputto the Buffer Driver). Moreover, note that the designation ‘-G’ denotesthe inverting aspect of the disclosed preferred embodiment of the BufferDriver. It would be obvious to one skilled in the art to utilize avariation of the disclosed Buffer Driver, that is non-inverting, coupledwith an Amplifier (that provides the gain stage) whose inputs arere-arranged in order to keep the signal signs in proper order in makingthe Buffer Amplifier.

As such, the amplifier gain stage's output node 414A (vo1_(400A)) ofFIG. 4A can connect to the buffer drivers vin1's, which can be the inputnode 414B of BUF_(400B), or input node 414C of BUF_(400C), or inputnodes 414D of BUF_(400D). Accordingly, gain output node 415A(vo1_(400A)) of FIG. 4A can connect to buffer driver vin2's, which canbe the input node 415B of BUF_(400B), or input node 415C of BUF_(400C),or input nodes 415D of BUF_(400D), respectively. Also, note that eitherone or both of the inputs of (inverting) buffer drivers, can be currentinput terminals.

In the next sections of this disclosure, AMP_(400B), AMP_(400C), orAMP_(400D) are configured in unity gain. In unity gain configuration,the output of BUF_(400B) which is Vo_(400B), or output of BUF_(400C)which is Vo_(400C), or output of BUF_(400D) which is V_(400D) connect tothe negative input terminal node 412A or v_(in−) of the amplifier gainstage of FIG. 4A. To explain how an amplifier's gain stage interactswith an (inverting) buffer driver, first assume an AMP₄₀₀ which iscomprised of amplifier gain stage of FIG. 4A coupled with an ideal(inverting) buffer driver, and describe both the steady state andnon-steady state of amplifier gain stage of FIG. 4A. In steady stateconditions, and assuming no non-idealities or mismatch, a AMP₄₀₀configured in unity gain (node 412A or v_(in−) is connected to theoutput of an ideal inverting buffer driver) would have v_(in−)=v_(in+)(i.e., Δv_(in)=0). In the input stage of FIG. 4A, there is currentequilibrium with ID_(M409A)≈ID_(M411A)≈0.5×ID_(M407A)≈i andID_(M408A)≈ID_(M410A)≈0.5×ID_(M406A). In the upper and lower currentmirrors, there is also current equilibrium, withID_(M403A)≈ID_(M413A)≈ID_(M415A)≈2i andID_(M402A)≈ID_(M412A)≈ID_(M414A)≈2i. Accordingly,(ID_(M416A)≈ID_(M418)≈i)≈(ID_(M417A)≈ID_(M419)≈i). Second, assume AMP₄₀₀is also in unity gain configuration, but it is initially innon-steady-state condition, for example v_(in−)<v_(in+) (albeit a verysmall finite −Δv_(in)). Here,(ID_(M410A)≈ID_(M408A))<(ID_(M409A)≈ID_(M411A)) andID_(M406A)>ID_(M407A). At node 405A, the increased ID_(M406A) ismirrored via ID_(M403A) to increase (ID_(M413A)≈ID_(M415A)). At node404A, the decreased ID_(M406A) is mirrored via ID_(M402A) to decrease(ID_(M412A)≈ID_(M414A)). With less of (ID_(M410A)≈ID_(M408A)) flowinginto nodes 411A and 409A, there will be more of the increased(ID_(M413A)≈ID_(M415A)) available to flow through(ID_(M417A)≈ID_(M419A)). Conversely, with more of(ID_(M409A)≈ID_(M411A)) flowing into nodes 410A and 408A, there will beless of the decreased (ID_(M412A)≈ID_(M414A)) available to flow through(ID_(M416A)≈ID_(M418)). Hence, initially, in non-steady state conditionwhenv_(in−)<v_(in+)→(ID_(M416A)≈ID_(M418A))<(ID_(M417A)≈ID_(M419A))→initiallyboth node 414A (Vo1_(400A)) and node 415A (Vo2_(400A)) are pulled downtogether. Conversely, when v_(in−)>v_(in+), the opposite dynamics occurand initially both node 414A (Vo1_(400A)) and node 415A (Vo2_(400A)) arepulled up together. Nodes 414A and 415A (are a pair of high-gainhigh-impedance outputs of the amplifier's gain stage that connect toinputs to the inverting buffer drivers, vin1 and vin2 that) have thesame polarity, and move up and down together.

Next, considering the AMP_(400B), or AMP_(400C), or AMP_(400D). Notethat steady state conditions may be disturbed indirectly as well. Forexample, a transitory current imbalance, Δi, at one of the current inputnode of the (inverting) buffer drivers that may be initiated bysink-source current in output FETs. In the presence of such transitoryΔi, the closed loop containing the amplifier gain stage (FIG. 4A) inconcert with BUF_(400B), or BUF_(400C), or BUF_(400D) could aim toregulate out the Δi until current balance is returned in steady state(e.g., by adjusting the voltages at node 414A and 415A viaID_(M416A)≈ID_(M418) and ID_(M417A)≈ID_(M419A)). This aspect of thedescription of AMP₄₀₀ will be relevant to the disclosures in the nextsections.

It would be possible that the buffer driver methodology described inthis disclosure would operate with other alternative amplifiers gainstages, beside that of FIG. 4A. Alternative amplifier gain stages tothat of FIG. 4A could be NMOS input FCTA, or PMOS input FCTA, FCTA withRGC, FCTA with input stage that contains double PMOS with an NMOS levelshifter, FCTA with input stage that contains double PMOS with an NMOSlevel shifter, or other non-FCTA configurations, amongst others. In eachof the next sections, the AMP_(400B), or AMP_(400C), or AMP_(400D) areconfigured in unity gain, and BUF_(400B), or BUF_(400C), or BUF_(400D)are described at steady state while sinking and sourcing current for anexternal load (i.e., sink and source mode). Also, for each of(inverting) buffer drivers, the steady state mode is described with noload.

Section (XIII): Detailed Description of Buffer Driver Comprised ofComplementary Non-Inverting Current Mirrors (CNICM), Minimum CurrentSelector (MCS) and Inverting Current Mirror Amplifier (ICMA),Illustrated in FIG. 4B (4B)

FIG. 4B is a circuit schematic showing a buffer driver (BUF_(400B))(e.g., inverting buffer driver) comprising of CNICM_(400B) andMCS_(400B) coupled with ICMA_(400B). This section build on what wasalready described in prior section. Note that the difference between theBUF_(400D) embodiment illustrated in FIG. 4D (which is described in itsrespective section), compared with the BUF_(400B) embodiment illustratedhere in FIG. 4B is the inclusion of CNICM_(400B). As such, the detaileddescription, including benefits are equally applicable between toBUF_(400D) of FIG. 4D and BUF_(400B) of FIG. 4B.

A buffer driver may be required to sink or source large amounts ofcurrent when interfacing with an external load. It would be advantageousto reduce the current consumption, in a buffer driver, that isattributed to the function of monitoring and sensing of output FETdriver sink-source currents (i.e. M_(438B), M_(435B)) that variesdepending on the size an external load. One of the benefits ofCNICM_(400B) here is to help reduce the buffer driver's load-dependentcurrent consumption. Also, CNICM_(400B) helps the buffer driver withhaving more consistent performance to specifications that are lessdependent on the external load variations.

Throughout the description of FIG. 4B, MOSFET scale factors (e.g., ‘s’,‘k’, or W/L rations for M_(421B)-M_(435B),M_(428B)-M_(429B)-M_(431B)-M_(433B), M_(434B)-M_(436B), etc.) andcurrent source scale factors (e.g., forI_(405B)-I_(406B)-I_(407B)-I_(408B)) can be modified depending oncost-performance goals such as die size, current consumption, speed, andcross over distortion, and others. For the FIG. 4B disclosure, s=k=20but ‘s’, ‘k’ can have wide ranges 0.01≤s≤1000000, 0.01≤k≤1000000. Theconnections of the elements in FIG. 4B are described as follows. Thebody terminal of all NMOSFETs in FIG. 4B are connected to node 2 whichis V_(SS), and the body terminals of PMOSFETs are connected to node 1which is V_(DD). For clarification, note that having one current source(e.g., I_(405B) and I_(406B)) with neither the upper nor the lowerterminal connected to a power supply (i.e., V_(DD), V_(SS)), is theequivalent of having two current sources where the first one has one ofits terminals connected to V_(DD) and the second one has one of itsterminals connected to V_(SS). Bias current sources I_(407B), andI_(408B) have their lower terminals connected to node 2, which isV_(SS). The source terminal of NMOSFETs M_(421B) M_(423B), M_(425B),M_(427B), M_(429B), M_(431B), and M_(435B) are connected to node 2 thatis V_(SS). The source terminal of PMOSFETs M_(420B), M_(422B), M_(424B),M_(426B), M_(428B), M_(430B), M_(432B), M_(434B), M_(436B), and M_(438B)are connected to node 1 that is V_(DD). Node 414B is the vin1_(400B)terminal of the buffer driver, and is connected to the gate terminal ofM_(420B) and the gate terminal of M_(438B). Node 415B is the vin2_(400B)terminal of the buffer driver, and is connected to the gate terminal ofM_(421B), the gate terminal of M_(435B), the drain terminal of M_(436B),and the upper terminal of current source I_(408B). Node 416B isconnected to the drain terminal of M_(420B), the drain terminal ofM_(422B), the gate terminal of M_(422B), the gate terminal of M_(424B),and the upper terminal of current source I_(405B). Node 417B isconnected to the drain terminal of M_(421B), the drain terminal ofM_(423B), the gate terminal of M_(423B), the gate terminal of M_(425B),and the lower terminal of current source I_(405B). Node 418B isconnected to the drain terminal of M_(424B), the drain terminal ofM_(426B), the gate terminal of M_(426B), the gate terminal of M_(428B),and the upper terminal of current source I_(406B). Node 419B isconnected to the drain terminal of M_(425B), the drain terminal ofM_(427B), the gate terminal of M_(427B), the gate terminal of M_(433B),and the lower terminal of current source I_(406B). Node 420B isconnected to the gate terminal of M_(429B), the drain terminal ofM_(429B), the drain terminal of M_(428B), and the gate terminal ofM_(431B). Node 421B is connected to the drain terminal of M_(431B) andthe source terminal of M_(433B). Node 422B is connected to the drainterminal of M_(433B), the gate terminal of M_(430B), the drain terminalof M_(430B), and the gate terminal of M_(432B). Node 423B is connectedto the drain terminal of M_(432B), the gate terminal of M_(434B), thedrain terminal of M_(434B), and the gate terminal of M_(436B), and theupper terminal of current source I_(407B). Node 424B is the Vo_(400B)terminal of the (inverting) buffer driver, and is connected to the drainterminal of M_(438B) and the drain terminal of M_(435B).

The (inverting) buffer driver or BUF_(400B) is comprised of BLOCK_(400B)which is the output FETs, or FET_(400B) (containing source PMOSFETM_(438B) and sink NMOSFET M_(435B)), plus BLOCK_(401B) which is theinverting current mirror amplifier or ICMA_(400B)(containing PMOSFETsM_(430B), M_(432B), M_(434B), M_(436B), and current sources I_(407B)I_(408B)), plus BLOCK_(402B) which is the minimum current selector orMCS_(400B) (containing PMOSFET M_(428B), and NMOSFETs M_(429B),M_(431B), and M_(433B)), plus BLOCK_(403B) which is the complementarynon-inverting current mirror or CNICM_(400B) (containing on PMOSFETsside M_(420B), M_(422B), M_(424B), M_(426B), and on the NMOSFET sideM_(421B), M_(423B), M_(425B), M_(427B), and current sources I_(405B) andI_(406B)). As noted earlier I_(405B) can be is the functional equivalentof 2 current sources, and is shown as one current source fordemonstrative simplicity. There can be I_(405B1) and I_(405B2) whereupper terminal of I_(405B1) is connected to V_(DD) and lower terminal ofI_(405B2) is connected to V_(SS). As such, lower terminal of I_(405B1)and upper terminal of I_(405B2) can be the functional equivalence of thelower and upper terminals of I_(405B). As stated earlier, AMP_(400B) iscomprised of the amplifier gain stage of FIG. 4A coupled withBUF_(400B), where node 414A is connected to node 414B, node 415A isconnected to node 415B. AMP_(400B) is configured in unity gain with node412A (the amplifier gain stage negative input terminal) connected tonode 420B (or Vo_(400B)).

Note that it would be possible to utilize a ICMA that is a complementaryversion of ICMA_(400B) (i.e., PMOSFET based or NMOSFET based). It wouldalso be possible to utilize an MCS that is a complementary version ofMCS_(400B) (i.e., PMOSFET base or NMOSFETs based). It would also bepossible that a plurality of MCS coupled with a plurality of MCSs can beutilized, which are complementary to one another (e.g., NMOSFET basedICMA coupled with primarily PMOSFET based MCS, utilized along withPMOSFET based ICMA coupled with primarily NMOSFET based MCS).

Moreover, it would be possible to design a buffer driver that utilizeother equivalent functional implementations that are complementarynon-inverting current mirrors (CNICM), minimum current selector (MCS) orinverting current mirror amplifier (ICMA). For example, a loser take all(LTA) can perform the equivalent function of MCS. Another example wouldbe to utilize, a current mirror amplifier (CMA) instead of annon-inverting current mirror amplifier (ICMA) by arranging the propercircuit signal signs. Another example would be to utilize a currentamplifier to perform the equivalent function of a ICMA.

In describing the function of CNICM_(400B), the non-idealities (e.g.,offsets and mismatches) are set aside and the boundary conditions arediscussed (when M_(438B) is maximally on and M_(435B) is off, and whenM_(438B) is off on and M_(435B) is maximally on). After that,CNICM_(400B)'s steady state condition is described (when the quiescentcurrents in M_(438B) and M_(435B) is equal to i×s, where MOSFET scalefactors s=k). The M_(438B) current is scaled down (by a factor of ‘s’)and mirrored by M_(420B). For clarity of description here, theID_(M420B) is the input, and ID_(M426B) is the output for PMOSFET sideof CNICM_(400B). If ID_(M438B)≈0→ID_(M420B)≈0→all of I_(405B)≈2i flowsonto ID_(M422B)≈2i→ID_(M424B)≈2i. Thus, all of I_(406B)≈2i is taken byM_(424B)→ID_(M426B)≈0, which in effect is the (zero scale) non-invertingcurrent mirror of ID_(M438B)≈0. IfID_(M438B)≈i_(max)→ID_(M420B)≈i_(max)×1/s. Assuming that i_(max)×1/s>2i.Note here that the maximum current consumed in ID_(M420B) is ‘2i’ thatsubstantially lowers V_(DS) of M_(420B), which substantially curbs theload-dependent source current consumption of BUF_(400B). Thus, all ofI_(405B)≈2i is taken by M_(420B)→ID_(M422B)≈0→ID_(M424B)≈0. Thus, all ofI_(406B)≈2i flows into M_(426B)→ID_(M426B)≈2i, which in effect is therectified non-inverting current mirror of ID_(M438B)≈i_(max). Note thatthe maximum (full scale) output current possible for M_(426B) is 2iwhich corresponds to i_(max) of M_(438B) (and its scaled down andmirrored current i_(max)×1/s of M_(420B)). Also note that ID_(M426B)that is monitored by MCS_(400B) (via M_(428B) M_(429B), M_(431B)) hasits full scale current capped at 2i enabling it to respond moreconsistently to i_(max) swings imposed on FET_(400B) for sourcingcurrent, for example, for heavy external loads (e.g., i_(max)>>2i×s).

For steady state, let's assume that there is no external load, and thatID_(M438B)≈s×i. Thus, ID_(M420B)≈i→the I_(405B)≈2i is split in half andID_(M420B)≈ID_(M422B)≈i→ID_(M424B)≈i. Thus, I_(406B)≈2i is also split inhalf→ID_(M424B)≈ID_(M426B)≈i→ID_(M426B)≈i, which in effect is the scaleddown (by 1/s) non-inverting current mirror of ID_(M438B)≈s×i. Similarsignal flow applies to the NMOSFET side of CNICM_(400B), where theM_(435B) current is scaled down (by a factor of ‘k’) and mirrored ontoM_(421B). If ID_(M435B)≈0→ID_(M421B)≈0→all of I_(405B)≈2i flows ontoID_(M423B)≈2i→ID_(M425B)≈2i. Thus, all of I_(406B)≈2i is taken byM_(425B)→ID_(M427B)≈0, which in effect is the (zero scale) non-invertingcurrent mirror of ID_(M435B)≈0. If ID_(M435B)≈i_(max)≈ID_(M421B)i_(max)×1/k. Let's assume that i_(max)×1/k>2i. Note here that themaximum current consumed in ID_(M421B) is ‘2i’ that substantially lowersV_(DS) of M_(421B), which substantially curbs the load-dependent sinkcurrent consumption of BUF_(400B). Thus, all of I_(405B)≈2i is taken byM_(421B)→ID_(M423B)≈0→ID_(M425B)≈0. Thus, all of I_(406B)≈2i flows intoM_(427B)→ID_(M427B)≈2i, which in effect is the rectified non-invertingcurrent mirror of ID_(M435B)≈i_(max). Note that the maximum (full scale)output current possible for M_(427B) is 2i which corresponds to i_(max)of M_(435B) (and its scaled down and mirrored current i_(max)×1/k ofM_(421B)). Also note that ID_(M427B) that is monitored by MCS_(400B)(via M_(433B)) has its full scale current capped at 2i enabling it torespond more consistently to i_(max) swings imposed on FET_(400B) forsinking current, for example, for heavy external loads (e.g.,i_(max)>>2i×k). For steady state, again it is assumed that there is noexternal load, and that ID_(M435B)≈k×i. Thus, ID_(M421B)≈i→theI_(405B)≈2i is split in half and ID_(M421B)≈ID_(M423B)≈i→ID_(M425B)≈i.Thus, I_(406B)≈2i is also split inhalf→ID_(M425B)≈ID_(M427B)≈i→ID_(M427B)≈i, which in effect is the scaleddown (by 1/k) non-inverting current mirror of ID_(M435B)≈k×i.

In summary, BLOCK_(403B) is CNICM_(400B) which is the complementarynon-inverting current mirror. One of the functions of CNICM_(400B) is tomonitor and mirror the currents in FET_(400B), and rectify the fullscale (maximal) mirrored currents that feed the MCS_(400B). Given thatthe sink-source currents of FET_(400B) can vary significantly, dependingon the external load, the scaled down but un-rectified mirroring ofFET_(400B) raw currents can also vary a lot, and increase the currentconsumption of a (inverting) buffer driver. Mirroring and rectifying thesink-source currents of FET_(400B) via CNICM_(400B), caps the full scaleand curbs the maximal mirrored currents (e.g., to ‘2i’) before thesink-source current signals are fed to MCS_(400B) for signal processing.

Therefore, arranging BUF_(400B) comprising CNICM_(400B) coupled withMCS_(400B) coupled with ICMA_(400B) provides the BUF_(400B) with addedbenefit of lower (load-dependent) sink-source current consumption, aswell as more consistent DC, AC, and transient performance, especiallyunder varying external load conditions. As stated earlier, descriptionof BUF_(400B) is provided considering that it is part of AMP_(400B) thatis configured in unity gain.

Before describing the BUF_(400B) operating in steady state and overdrive (sink-source), the current equilibrium conditions are described atnodes 414A, 414B and 415A, 415B for steady state to persist. At nod 415Bin FIG. 4B (which is connected to node 415A in FIG. 4A), KCL requiresthe sum of ID_(M416A), ID_(M417A), I_(408B), and ID_(M436B) to convergeto zero in order for steady state to hold. Also, at node 414B in FIG. 4B(which is connected to node 414A in FIG. 4A), KCL requires the sum ofID_(M418A), and ID_(M419A) to converge to zero for steady state to hold.Moreover, for AMP₄₀₀ unity gain loop (containing amplifier gain stage ofFIG. 4A and BUF_(400B)) to stay in steady state,(ID_(M417A)≈ID_(M419A))≈(ID_(M416A)≈ID_(M418A)) and I_(408B)≈ID_(M429B).This is called “the conditions of current equilibrium for nodes 414B and415B” which will be referred to below.

During steady-state conditions, ID_(M421B)≈ID_(M427B)≈i (with k≈s, andcorresponding to ID_(M435B)≈k×i), and ID_(M420B)≈ID_(M426B)≈i (with k≈s,and corresponding to ID_(M438B)≈s×i). Therefore,ID_(M426B)≈ID_(M428B)≈ID_(M429B)≈ID_(M431B)≈i→VGS_(M427B)≈VGS_(M429B)≈VGS_(M431B).As explained in the previous sections above, with M_(429B) and M_(427B)(both with W/L=1×) operating at current ‘i’ while providing equal gatevoltages for M_(431B) and M_(433B) that are in series (both withW/L=2×), causes the operating current of M_(431B) and M_(433B) to alsobe ‘i’. As such, the MCS_(400B) selects current equality whenmin(M_(426B), M_(427B))≈min(M_(428B), M_(433B))≈M_(433B)≈M_(428B)≈i. TheID_(M433B)≈i is the output of MCS_(400B) that is fed onto M_(430B) whichis the input to ICMA_(400B). Here, for the current mirrorID_(M430B)≈i≈ID_(M432B)→the current source I_(407B)≈2i is split equallybetween ID_(M432B)≈i≈ID_(M434B)→the current mirrorID_(M434B)≈ID_(M436B)≈i. The ID_(M436B)≈i is in balance with the currentsource I_(408B)=i. This is how current equilibrium is held at node 415B,which sustains the controlled quiescent currents in the buffer driveroutput FETs, ID_(M438B)≈ID_(M435B)≈s×i (assume setting k≈s).

A brief summary is provided here for BUF_(400B) (contained inAMP_(400B)) of FIG. 4B, utilizing CNICM_(400B) coupled with MCS_(400B)coupled with ICMA_(400B), for the sink and source modes when there islarge difference between ID_(M438B) and ID_(M435B). As a reminder, k=sand ID_(M435B) is rectified by CNICM_(400B) and presented to MCS_(400B)via ID_(M427B), and ID_(M438B) is rectified by CNICM_(400B) andpresented to MCS_(400B) via ID_(M426B).

In the sink mode, node 414B and node 415B are pulled up towards V_(DD),MN_(435B) and MN_(421B) are turned on→ID_(M421B)>>i→initially beforecurrent equilibrium ID_(M427B)≈2i. Also, MN_(438B) and MN_(420B) areturned off or are nearly off,initially→ID_(M426B)≈ID_(M428B)≈ID_(M429B)≈ID_(M431B)≈0 which wouldblock current flow in ID_(M433B)→MCS_(400B) selects the min(ID_(M427B),ID_(M426B))=ID_(M426B)≈0<<i, which starves M_(433B), andinitially→ID_(M432B)≈ID_(M430B)<<i→all of I_(407B)=2i flows intoM_(434B)→ID_(M434B)≈ID_(M436B)≈2i at node 415B with I_(408B)=i, which ina current imbalance at node 415B initially, and as such the AMP_(400B)loop kicks to bring back the balance. In order for “conditions ofcurrent equilibrium for nodes 414B and 415B” to return, and to simplifythe explanation for clarity of description, in FIG. 4A(ID_(M417A)≈ID_(M419A)) can get very slightly increased above(ID_(M416A)≈ID_(M418A)). Here, by AMP_(400B) loop regulation, thevoltages at both node 414B and node 415B are pulled down sufficientlyenough until M_(420B) (and its mirror M_(438B)) turn back on enough forID_(M420B)ID_(M426B)≈i→ID_(M428B)≈ID_(M429B)≈ID_(M431B)≈i→by movingtowards current balance, MCS_(400B) inputs are regulated and as suchmin(ID_(M426B), ID_(M427B))≈ID_(M426B)≈i. Such is the case sinceM_(435B) is still sinking current,ID_(M435B)>ID_(M438B)→ID_(M421B)>ID_(M420B)→ID_(M427B)>ID_(M426B). Asexplained in the previous section, note that in FIG. 4A, despite thevery slight increase in (ID_(M417A)≈ID_(M419A)) above(ID_(M416A)≈ID_(M418A)) to regulate down the voltages at node 414B and415B, ICMA_(400B) (coupled and being fed via MCS_(400B) andCNICM_(400B)) have enough gain to regulate a slightly increasedID_(M436B) (as compared to the fixed I_(408B)≈i) to keep the voltage atnode 415B high enough so that M_(435B) can continue sinking highcurrents to the external load. Here, the “conditions of currentequilibrium for nodes 414B and 415B” are held, and steady stateconditions maintained, while M_(426D) to continue sourcing current.Accordingly, ID_(M420B)≈i→the operating current in the inactivesource-FET is thus controlled and regulated at ID_(M431B)≈s×i, whilesteady state condition is maintained and the sink FET, M_(435B),continues sinking extra current to an external load.

Conversely, in the source mode, when node 414B and node 415B of FIG. 4Bare pushed down towards V_(SS), initially M_(438B) and M_(420B) turn onand M_(435B) and M_(421B) turn off→then initially before equilibrium,ID_(M426B)≈ID_(M428B)≈ID_(M429B)≈ID_(M431B)≈2i. Moreover,ID_(M427B)≈0→VG_(M427B)=0=VG_(M433B)→M_(433B) blocks flow ofID_(M431B)→thus, before equilibrium, MCS_(400B) selects the min(ID_(M427B), ID_(M426B))≈ID_(M426B)≈0→output of MCS_(400B) isID_(M433B)≈0 that is fed onto M_(430B) which is the input ofICMA_(400B). ID_(M430B)≈0≈ID_(M432B)→all of current in I_(407B)≈2i,before equilibrium, could flows in the current mirrorID_(M434B)≈ID_(M436B)≈2i>I_(408B)≈1i, which creates a current imbalanceat node 415B that is unsustainable and as such ICMA_(400B) containedwithin the AMP_(400D) loop kicks in to regulate back towards currentequilibrium. In order for “conditions of current equilibrium for nodes414B and 415D” to hold, and to simplify the explanation for clarity ofdescription, (ID_(M416A)≈ID_(M418A)) (ID_(M417A)≈ID_(M419A)).Concurrently, voltage at node 415B gets regulated by being pulled up byID_(M436B) until ID_(M436B)≈i=I_(408B). In order to arrive at thiscurrent equilibrium, ICMA_(400B) along with AMP_(400B) loop regulate thevoltage of node 415B down sufficiently enough untilID_(M436B)≈i=I_(408B)→ID_(M421B)≈ID_(M427B)≈i. By moving towards currentbalance, MCS_(400B) inputs are regulated and as such min (ID_(M426B),ID_(M427B))≈ID_(M427B)≈i. Such is the case because M_(438B) is stillsourcing high currents to for an externalload→ID_(M435B)<ID_(M438B)→ID_(M421B)<ID_(M420B)→ID_(M427B)<ID_(M426B).As such, when the buffer driver sources current for an output load, thenM_(435B) which is the inactive sink FET, can operate at a controlledregulated current ID_(M435B)≈k×i. Again, note that ‘i’ is a constantcurrent source set by the main bias network of the amplifier, and ‘k’ isFET aspect ratio that is tightly controllable, which helps with thecontrol of the quiescent current in the inactive sink FET.

In summary, when there is an imbalance between sink-source FETscurrents, ID_(M438B) and ID_(M435B), such imbalance is scaled down,mirrored, and rectified by the complementary non-inverting currentmirror (CNICM_(400B)). The outputs of CNICM_(400B), ID_(M427B) andID_(M426B), which are the rectified version of sink-source currents, arethen fed onto the MCS_(400B). The MCS_(400B) effectively selects the(scaled) minimum current between the output drive currents ID_(M438B)and ID_(M435B). The selected minimum current, at the output ofMCS_(400B), is then inputted onto ICMA_(400B) which, together with theoperations of amplifier's gain stage, regulates a minimum quiescentcurrent in the inactive FET (e.g. for s=k, when ID_(M435B) is sinkingcurrent for an external load→inactive source FET ID_(M438B)=s×i, andwhen ID_(M438B) is sourcing current for an external load→inactive sinkFET ID_(M435B)=k×i).

In conclusion, as noted in the previous section, the benefits of theFIG. 4D's BUF_(400D) are equally applicable to that of FIG. 4B'sBUF_(400B) (comprising CNICM couple with MCS coupled with ICMA) whichare briefly summarized here again. First, the function of ‘monitoring’,of sink-source currents, consume current itself. Sink-source currentscan be high and have unpredictable patterns. The CNICM effectively curbs(e.g., rectify) the ‘monitoring’ current consumption. Also therectifying of output FET currents, before they are fed to the MCS andICMA, helps with consistency of DC, AC, and transient performance of thebuffer driver, with more independence from external load variations.Second, the buffer driver is fast since the CNICM couple with MCScoupled with ICMA are all in principal operating in current mode, whichis inherently fast. Third, it can operate with low power supply voltagesince the minimum V_(DD) is V_(GS)+2V_(DS) (low voltage) for CNICM, MCS,and ICMA. Fourth, it is made of a few transistor which makes it smalland low cost. Fifth, buffer driver can operate with ultra low currentand ultra low power, and utilization of CNICM helps curb the currentconsumption attributed with monitoring the sink-source currents when theamplifier drives different external loads. Sixth, it has wideinput-output span of near rail-to-rail. Seventh, it can provide highsink-source drive capability and controlled quiescent current in theinactive output FET that benefits DC and dynamic performance. FIG. 4E isa simulation of a buffer amplifier containing buffer driver of FIG. 4Bcoupled with gain stage of FIG. 4A. This simulations indicatesapproximate and typical sink-source current capability for the bufferamplifier to drive external load resistors 900KΩ, 300KΩ, and 30KΩ. Thesimulation indicates current consumption for the buffer amplifier ofroughly 120 nA for FIG. 4B coupled with gain stage of FIG. 4A. Eighth,buffer driver is minimally imposing on the amplifier's gain or the speedof the preceding amplifier's high-gain stage Ninth, it's based onstandard CMOS process which is low cost, high quality, ready available).Tenth, the buffer driver arrangement utilizing CNICM coupled with MCScoupled with ICMA can be tailored to very high current and very highspeed.

Section (XIV): Detailed Description of Buffer Driver Comprised ofComplementary Non-Inverting Current Mirrors (CNICM), Loser Take all(LTA) and Current Mirror Amplifier (CMA), Illustrated in FIG. 4C (4C)

FIG. 4C is a circuit schematic showing a (inverting) buffer driver(BUF_(400C)) comprising of CNICM_(400C) and LTA_(400C) coupled withCMA_(400C). This section builds on what was already described inprevious section. A difference between the BUF_(400C) embodimentillustrated in FIG. 4B in the previous section compared to theBUF_(400C) embodiment illustrated in FIG. 4C is the utilization ofLTA_(400C) which is functionally equivalent to MCS_(400B). As such, thedetailed description (e.g., regarding AMP_(400B), CNICM_(400B),CMA_(400B)) that were provided in prior section pertaining to BUF_(400B)of FIG. 4B are equally applicable to that of BUF_(400C) of FIG. 4C.

Throughout the description of FIG. 4C, MOSFET scale factors (e.g., ‘s’,‘k’, or W/L rations for M_(421C)-M_(439C), M_(432C)-M_(420C),M_(426C)-M_(428C), M_(427C)-M_(429C), etc.) and current source scalefactors (e.g., for I_(405C)-I_(406C)-I_(407C)) can be modified dependingon cost-performance goals such as die size, current consumption, speed,and cross over distortion, and others. For example, the FIG. 4Cdisclosure, s=k=20 but ‘s’, ‘k’ can have wide ranges 0.01≤s≤1000000,0.01≤k≤1000000.

The connections of the elements in FIG. 4C are described as follows. Thebody terminal of all NMOSFETs in FIG. 4C are connected to node 2 whichis V_(SS), and the body terminals of PMOSFETs are connected to node 1which is V_(DD). Bias current sources I_(407C) upper terminals connectedto node 1, which is V_(DD). The source terminal of NMOSFETs M_(421C)M_(423C), M_(425C), M_(427C), M_(429C), M_(431C), M_(433C), M_(435C),M_(437C), and M_(439C) are connected to node 2 that is V_(SS). Thesource terminal of PMOSFETs M_(420C), M_(422C), M_(424C), M_(426C),M_(428C), M_(430C), and M_(432C) are connected to node 1 that is V_(DD).Node 414C is the vin1_(400C) terminal of the buffer driver, and isconnected to the gate terminal of M_(420C) and the gate terminal ofM_(432C). Node 415C is the vin2_(400C) terminal of the buffer driver,and is connected to the gate terminal of M_(421C), the gate terminal ofM_(439C), the drain terminal of M_(437C), and the lower terminal ofcurrent source I_(407C). Node 416C is connected to the drain terminal ofM_(420C), the drain terminal of M_(422C), the gate terminal of M_(422C),the gate terminal of M_(424C), and the upper terminal of current sourceI_(405C). Node 417C is connected to the drain terminal of M_(421C), thedrain terminal of M_(423C), the gate terminal of M_(423C), the gateterminal of M_(425C), and the lower terminal of current source I_(405C).Node 418C is connected to the drain terminal of M_(424C), the drainterminal of M_(426C), the gate terminal of M_(426C), the gate terminalof M_(428C), the gate terminal of M_(430C), and the upper terminal ofcurrent source I_(406C). Node 419C is connected to the drain terminal ofM_(425C), the drain terminal of M_(427C), the gate terminal of M_(427C),the gate terminal of M_(429C), and the lower terminal of current sourceI_(406C). Node 420C is connected to the drain terminal of M_(429C), thedrain terminal of M_(428C), the drain terminal of M_(431C), the gateterminal of M_(431C), and the gate terminal of M_(433C). Node 421C isconnected to the drain terminal of M_(430C), the drain terminal ofM_(433C), the gate terminal of M_(435C), the drain terminal of M_(435C),and the gate terminal of M_(437C). Node 422C is the Vo_(400C) terminalof the (inverting) buffer driver, and is connected to the drain terminalof M_(432C) and the drain terminal of M_(439C).

The buffer driver (e.g., inverting buffer drive) or BUF_(400C) iscomprised of BLOCK_(400C) which is the output FETs, or FET_(400C)(containing source PMOSFET M_(432C) and sink NMOSFET M_(439C)); plusBLOCK_(401C) which is the inverting current mirror amplifier orCMA_(400C) (containing NMOSFETs M_(435C), and M_(437C), and currentsources I_(407C)); plus BLOCK_(402C) which is the loser take all orLTA_(400C) (containing PMOSFET M_(428C), and M_(430C), and NMOSFETsM_(429C), M_(431C), and M_(433C)); plus BLOCK_(403C) which is thecomplementary non-inverting current mirror or CNICM_(400C) (containingon PMOSFETs side M_(420C), M_(422C), M_(424C), M_(426C), and on theNMOSFET side M_(421C), M_(423C), M_(425C), M_(427C), and current sourcesI_(405C) and I_(406C)). As stated earlier, AMP_(400C) includes anamplifier gain stage of FIG. 4A coupled with BUF_(400C), where node 414Ais connected to node 414C, node 415A is connected to node 415C.AMP_(400C) is configured in unity gain with node 412A (the amplifiergain stage negative input terminal) connected to node 422C (orVo_(400C)). Note that it would be possible to utilize a CMA that iscomplementary version of CMA_(400C) (i.e., PMOSFET based or NMOSFETbased). Also, it would also be possible to utilize an LTA that is acomplementary version of LTA_(400C) (i.e., primarily PMOSFET base orNMOSFETs based). It would also be possible that plurality of CMAscoupled with plurality of LTAs can be utilized, which are complementaryto one another (e.g., NMOSFET based CMA coupled with primarily NMOSFETbased LTA, utilized along with PMOSFET based CMA coupled with primarilyPMOSFET based LTA). Moreover, it would be possible to design a bufferdriver that utilizes other equivalent functional implementations ofCNICM, LTA or CMA. For example, LTA can perform the equivalent functionof MCS. Another example would be to utilize, current mirror amplifier(CMA) instead of CMA by properly arranging the circuit signal signs.This can be done, for example, by arranging an inverting LTA feeding aCMA, instead of LTA feeding a CMA. Another example would be to utilize acurrent amplifier (CA) or an inverting CA to perform the equivalentfunction of a CMA or CMA, respectively.

The detailed description of the CNICM_(400C), LTA_(400C), CMA_(400C)were provided in previous sections, and their inner working as part ofAMP_(400C) loop will be be briefly discussed in this section. Forclarity of description, non-idealities (e.g., offsets or mismatches) areset aside in these discussions. Before describing the BUF_(400C)operating in steady state conditions and over drive (sink-source), thecurrent equilibrium conditions are described at nodes 414A, 414C and415A, 415C in order for the steady state conditions to persist. At node415C in FIG. 4C (which is connected to node 415A in FIG. 4A), KCLrequires the sum of ID_(M416A), ID_(M417A), I_(408C), and ID_(M436C) toconverge to zero. Also, at nod 414C in FIG. 4C (which is connected tonode 414A in FIG. 4A), KCL requires the sum of ID_(M418A), andID_(M419A) to converge to zero. Moreover, for AMP₄₀₀ unity gain loop(containing amplifier gain stage of FIG. 4A and BUF_(40C)) to stay insteady state, (ID_(M417A)≈ID_(M419A))≈(ID_(M416A)≈ID_(M418A)) andI_(407C)≈ID_(M437C). This is called “the conditions of currentequilibrium for nodes 414C and 415C” in this section, which will bereferred to herein this description.

As explained in the previous section, BLOCK_(403C) is CNICM_(400C) whichis the complementary non-inverting current mirror, and one of itsfunction is to monitor by mirroring the FET_(400C) currents and rectifythem before feeding the LTA_(400C) for signal processing. Mirroring andrectifying the sink-source currents of FET_(400C) via CNICM_(400C),curbs the maximal mirrored currents (e.g., to ‘2i’) before they are fedto LTA_(400C). Setting k=s, for steady state conditions (with noexternal load), it is assumed thatID_(M439C)≈k×i≈ID_(M432C)≈s×i≈ID_(M421C)≈i≈ID_(M420C)→the I_(405C)≈2i issplit in half→ID_(M421C)≈ID_(M423C)≈i≈ID_(M425C) andID_(M420C)≈ID_(M422C)≈i≈ID_(M424C)→the I_(406C)≈2i is split inhalf→ID_(M424C)≈i≈ID_(M426C)≈ID_(M428C)≈ID_(M430C), and ID_(M425C)i≈ID_(M427C)≈ID_(M429C). The ID_(M426C) and ID_(M427C) represent theinputs and ID_(M435C) represents the output of LTA_(400C). As indicatedbefore, one of the tasks of minimum current selector (MCS) or itsfunctional equivalent LTA_(400C) (in the embodiment of BUF_(400C)) hereis to monitor the sink-source signals, in this case through CNICM_(400C)that provides ID_(M426C) and ID_(M427C) which are then fed ontoLTA_(400C) that outputs the current ID_(M435C). In steady state, withID_(M428C)≈i≈ID_(M429C)→at node 420C,ID_(M428C)−ID_(M429C)≈ID_(M431C)≈0≈ID_(M433C) there is no current leftfor M_(431C)→ID_(M431C)≈0≈ID_(M433C)→at node 421C, with ID_(M430C)≈i,and ID_(M430C)−ID_(M433C)≈ID_(M435C)≈i. In summary, in steady state,LTA_(400C) responds to current equality with LTA (ID_(M426C),ID_(M427C))≈ID_(M426C)≈ID_(M427C)≈i, which is fed onto the input ofICMA_(400C) via M_(435C). The functions of the CMA_(400C), as containedin the AMP_(400C), is to help regulate the steady-state current inFET_(400C). Accordingly, ID_(M435C)≈i→ID_(M437C)≈i≈I_(407C). Here,AMP_(400C) loop is in steady-state, and node 414C and 415C currents arebalanced with (ID_(M417A)≈ID_(M419A))≈(ID_(M416A)≈ID_(M418A)) andI_(407C)≈ID_(M437C).

Next will be described the sink conditions. In the sink mode, node 414Cand node 415C are pulled up towards V_(DD), MN_(439C) and MN_(421C) areturned on→ID_(M421C)>>i→initially before current equilibriumID_(M427C)≈2i, which feeds one of LTA_(400C) inputs. Also, M_(432C) andM_(420C) are turned off or are nearly off, initially→ID_(M426C)≈0 whichfeeds the other input of LTA_(400C)→ID_(M426C)≈ID_(M428C)≈ID_(M430C)≈0and ID_(M427C)≈2i≈ID_(M429C), which pulls down on the voltage at node420C towards V_(SS)→ID_(M431C)≈0≈ID_(M433C)→withID_(M430C)≈0≈ID_(M433C), then ID_(M430C)−ID_(M433C)≈0 is the net currentoutput of the LTA_(400C) that is fed onto M_(435C), as the input ofCMA_(400C). In summary so far, before equilibrium, the LTA (ID_(M426C),ID_(M427C))≈ID_(M426C)≈0 that is fed onto CMA_(400C) viaM_(435C)→ID_(M435C)≈ID_(M437C)≈0<I_(407C)=i at node 415C, which is acurrent imbalance, initially→AMP_(400C) loop kicks to bring back thebalance. In order for “conditions of current equilibrium for nodes 414Cand 415C” to return here, and to simplify the explanation for clarity ofdescription, in FIG. 4A (ID_(M417A)≈ID_(M419A)) can increase slightlyover (ID_(M416A)≈ID_(M418A))→AMP_(400C) loop can regulate down V_(414C)and V_(415C) slightly enough until M_(420C) (and its mirror M_(432C))turn back on enough for ID_(M420C)≈ID_(M426C)≈i≈ID_(M428)≈ID_(M430C).Moving towards steady state while, MN_(439C) is sinking current for anexternalload→ID_(M421C)>>i→ID_(M427C)≈2i≈ID_(M429C)→ID_(M429C)≈2i>ID_(M428C)≈i→M_(429C)pulls down on the voltage at node 420C towardsV_(SS)→ID_(M431C)≈0≈ID_(M433C)→moving towards steady state, withID_(M430C)≈i, then ID_(M430C)−ID_(M433C)≈i is the net current output ofthe LTA_(400C) that is fed onto M_(435C), as the input of CMA_(400C).Note that LTA(ID_(M426C), ID_(M427))≈ID_(M426C)≈i. In summary, this isthe case for LTA_(400C) since M_(439C) is still sinking current,ID_(M439C)>ID_(M432D)→ID_(M421C)>ID_(M420C)→ID_(M427C)>ID_(M426C). Asexplained in the previous section, note that in FIG. 4A, despite thevery slight increase in (ID_(M417A)≈ID_(M419A)) over(ID_(M416A)≈ID_(M418A)) that can regulate down V_(414C) and V_(415C),then CMA_(400C) (coupled and being fed via LTA_(400C) and CNICM_(400C))have enough gain to regulate ID_(M437C) low enough (below the fixedI_(407C) i) in order V_(415C) to be high enough so that M_(439C) cancontinue sinking high currents to the external load. Here, the“conditions of current equilibrium for nodes 414C and 415C” are held,and steady state conditions maintained, while M_(439C) to continuesourcing current for an external load. Accordingly,ID_(M420C)≈i→operating current in the inactive source-FET is thuscontrolled and regulated at ID_(M431C)≈s×i, while steady state conditionis maintained and the sink FET, M_(439C), continues sinking extracurrent to an external load.

Conversely, in the source mode, when node 414C and node 415C of FIG. 4Care pushed down towards V_(SS), initially M_(432C) and M_(420C) turn onand M_(439C) and M_(421C) turn off→then initially before equilibrium,ID_(M426C)≈2i and ID_(M427C)≈0. In the LTA_(400C),ID_(M426C)≈2i≈ID_(M428C)≈ID_(M430C) andID_(M427C)≈0≈ID_(M429C)→ID_(M428C)−ID_(M429C)≈2i≈ID_(M431C)≈ID_(M433C)→beforeequilibrium, ID_(M430C)−ID_(M433C)≈2i−2i≈0 is the net current output ofthe LTA_(400C) that is fed onto M_(435C) as the input ofCMA_(400C)→ID_(M435C)≈0. In summary so far, before equilibrium, the LTA(ID_(M426C), ID_(M427C))≈ID_(M427C)≈0 that is fed ontoCMA_(400C)→ID_(M435C)≈ID_(M437C)≈0 at node 415C with I_(407C)=i, whichin a current imbalance at node 415C initially, that isunsustainable→CMA_(400C), contained within the AMP_(400C) loop, liftsV_(415C) enough until ID_(M437C)≈i_(407C)→‘conditions of currentequilibrium for nodes 414C and 415C’ can be maintained when(ID_(M416A)≈ID_(M418A))≈(ID_(M417A)≈ID_(M419A)) andID_(M437C)≈i=I_(407C), which occurs after AMP_(400C) loop regulatesV_(415C) high enough→ID_(M421C)≈ID_(M427C)≈i≈ID_(M429C), while MN_(432C)continues to source current for an externalload→ID_(M420)>>i→ID_(M426C)≈2i≈ID_(M428C)≈ID_(M430C)→ID_(M428C)−ID_(M429C)≈i,which is fed onto ID_(M431C)≈i≈ID_(M433C)→moving towards steady state,ID_(M430C)−ID_(M433C)≈i, which is fed onto input of CMA_(400C) that isID_(M435C)≈i≈ID_(M437C)≈I_(407C)≈i at node 415. In summary, note thatsince M_(432C) is still sourcing current, while in equilibriumID_(M432C)>ID_(M439C)→ID_(M420C)>ID_(M421C)→ID_(M426C)>ID_(M427C)→atequilibrium LTA(ID_(M426C), ID_(M427))≈ID_(M427C)≈i which is fed ontoCMA_(400C) via ID_(M435C)≈i ID_(M437C)≈I_(407C)≈i. As such, when thebuffer driver sources current for an output load, then M_(439C) which isthe inactive sink FET, can operate at a controlled regulated currentID_(M439C)≈k×i. Again, note that ‘i’ is a constant current source set bythe main bias network of the amplifier, and ‘k’ is FET aspect ratio thatis tightly controllable, which helps with the control of the quiescentcurrent in the inactive sink FET.

In summary, when there is an imbalance between sink-source FETscurrents, ID_(M432C) and ID_(M439C), such imbalance is scaled down,mirrored, and rectified by the complementary non-inverting currentmirror (CNICM_(400B)). The outputs of CNICM_(400C), ID_(M427C) andID_(M426C), which are the rectified version of sink-source currents, arethen fed onto the LTA_(400C). The LTA_(400C) effectively performs theequivalent function of selecting the (scaled) minimum current betweenthe output drive currents ID_(M432C) and ID_(M439C). The selectedminimum current, at the output of LTA_(400C), is then inputted ontoCMA_(400C) which, together with the operations of amplifier's gain stageinside the AMP_(400C) loop, regulates a minimum quiescent current in theinactive FET (e.g. for s=k, when ID_(M439C) is sinking current for anexternal load→inactive source FET ID_(M432C)=s×i, and when ID_(M432C) issourcing current for an external load→inactive sink FET ID_(M439C)=k×i).

In conclusion, as noted in the previous section, the benefits of theFIG. 4B's BUF_(400B) are equally applicable to that of FIG. 4C'sBUF_(400C) (comprising CNICM coupled with LTA coupled with CMA) whichare briefly summarized here again. First, the function of ‘monitoring’,of sink-source currents, consumes current itself. Sink-source currentscan be high and have unpredictable patterns. The CNICM effectively curbs(e.g., rectify) the ‘monitoring’ current consumption overhead. Also therectifying of output FET currents, before they are fed to the LTA andCMA, helps with consistency of DC, AC, and transient performance of thebuffer driver, with more independence from external load variations.Second, the buffer driver is fast since the CNICM couple with LTAcoupled with CMA are all in principal operate in current mode, which isinherently fast. Third, it can operate with low power supply voltagesince the minimum V_(DD) is V_(GS)+2V_(DS) (low voltage) for CNICM, LTA,and CMA. Fourth, it is made of a few transistor which makes it small andlow cost. Fifth, buffer driver can operate with ultra low current andult low power. The utilization of CNICM helps curb the currentconsumption attributed with monitoring the sink-source currents when theamplifier drives different external loads. Sixth, it has wideinput-output span of near rail-to-rail. Seventh, it can provide highsink-source drive capability and controlled quiescent current in theinactive output FET that benefits DC and dynamic performance. FIG. 4F isa simulation of a buffer amplifier containing buffer driver of FIG. 4Ccoupled with gain stage of FIG. 4A. This simulations indicatesapproximate and typical sink-source current capability for the bufferamplifier to drive external load resistors 900KΩ, 300KΩ, and 30KΩ. Thesimulation indicates current consumption for the buffer amplifier ofroughly 110 nA for FIG. 4C coupled with gain stage of FIG. 4A. Eighth,buffer driver is minimally imposing on the amplifier's gain or the speedof the preceding amplifier's high-gain stage. Ninth, it's based onstandard CMOS process which is low cost, high quality, ready available).Tenth, the buffer driver arrangement utilizing CNICM coupled with LTAcoupled with CMA can be tailored to very high current and very highspeed.

Section (XV): Detailed Description of Buffer Driver (BUF) Comprised ofMinimum Current Selector (MCS) and Inverting Current Mirror Amplifier(ICMA), Illustrated in FIG. 4D (4D)

FIG. 4D is a circuit schematic showing a buffer driver (e.g., invertingbuffer diver), BUF_(400D), comprising MCS_(400D) coupled withICMA_(400D). This last section is started by summarizing thedifferentiation of the last 2 sections, that is equally applicable tothis section's disclosure regarding buffer driver MCS_(400D) coupledwith ICMA_(400D): (1) can operate at low power supply voltage, withV_(DD) as low as V_(GS)+2V_(DS), (2) principally operates in currentmode which makes it fast and widens its dynamic range, (3) facilitatesnear rail-to-rail input-output operation, (4) it is simple and made offew transistors, making it inexpensive, (5) can sink and source largecurrents to heavy output loads, while keeping its internal currentconsumption ultra low (6) can control the inactive sink-source outputFET current, and thus improve its DC and dynamic performance (7) can runat ultra-low currents, (8) impose minimal loading back into thepreceding amplifier's high-impedance stage and as a result attainhigh-gain amplification, (9) impose minimal low frequency poles, thatmight otherwise burden the AC and transient response of the amplifier'sgain stage, and as a result improve an amplifier's dynamic response (10)can be made using standard CMOS process which is inexpensive and highquality, and (11) the advantages of this teaching are not exclusive toultra-low power, but this teaching can be tailored for buffer driversthat need very high-speed, very high drive current capability, andprovide near rail-to-rail input-outputs, with low supply voltages aswell.

While the buffer drivers sinks or source large currents for an externalload, for continued proper performance including fast dynamic response,low cross over distortion, and low output impedance, buffer driversgenerally run a minimum (well controlled) quiescent current in theirinactive output FET drivers. In this embodiment, the MCS_(400D) sensethe BUF_(400D)'s output FET driver's (M_(426D)-M_(431D)) current signalsand feed a proportional signal into ICMA_(400D). While the output FET issinking current, the MCS_(400D) coupled with ICMA_(400D), and in concertwith AMP_(400D) loop, regulate the minimum current in the (inactive)source output FET. Conversely, while the output FET is sourcing current,the MCS_(400D) coupled with ICMA_(400D), and in concert with AMP_(400D)loop, regulate the minimum current in the (inactive) sink output FET.

Throughout the description of FIG. 4D, MOSFET scale factors (e.g., ‘s’,‘k’, or W/L ratios for M_(421D)-M_(422D), M_(424D), M_(423D)-M_(425D),etc) and current source scale factors (e.g., for I_(407D)-I_(408D)) canbe altered depending on cost-performance goals such as die size, currentconsumption, speed, and cross over distortion, and other goals. For theFIG. 4D disclosure, s=k=20 but ‘s’, ‘k’ can have wide ranges0.01≤s≤1000000, 0.01≤k≤1000000. The connections of the elements in FIG.4D are described as follows. The body terminal of all NMOSFETs in FIG.4D are connected to node 2 which is V_(SS), and the body terminals ofPMOSFETs are connected to node 1 which is V_(DD). Bias current sourcesI_(407D) and I_(408D) have their upper terminals connected to node 1,which is V_(DD). The source terminal of NMOSFETs M_(421D), M_(423D),M_(425D), M_(427D), M_(429D), and M_(431D), are connected to node 2 thatis V_(SS). The source terminal of PMOSFETs M_(420D), M_(422D), andM_(426D) are connected to node 1 that is V_(DD). Node 414D is thevin1_(400D) terminal of the buffer driver, and is connected to the gateterminal of M_(424D), the gate terminal of M_(426D), the drain terminalof M_(429D), and the lower terminal of current source I_(408D). Node415D is the vin2_(400D) terminal of the buffer driver, and is connectedto the gate terminal of M_(421D) and the gate terminal of M_(431D). Node416D is connected to the gate terminal of M_(420D), the drain terminalof M_(420D), the drain terminal of M_(421D), and the gate terminal ofM_(422D). Node 417D is connected to the drain terminal of M_(422D) andthe source terminal of M_(424D). Node 418D is connected to the drainterminal of M_(424D), the gate terminal of M_(423D), the drain terminalof M_(423D), and the gate terminal of M_(425D). Node 419D is connectedto the drain terminal of M_(425D), the gate terminal of M_(427D), thedrain terminal of M_(427D), and the gate terminal of M_(429D), and thelower terminal of current source I_(407D). Node 420D is the VO_(400D)terminal, which is the output of the buffer driver, and is connected tothe drain terminal of M_(426D) and the drain terminal of M_(431D).

The buffer driver (e.g., inverting buffer driver) is comprised ofBLOCK_(400D) which is the output FETs, or FET_(400D) (containing sourcePMOSFET M_(426D) and sink NMOSFET M_(431D)), BLOCK_(401D) which is thenon-inverting current mirror amplifier or ICMA_(400D) (containingNMOSFETs M_(423D), M_(425D), M_(427D), M_(429D), and current sourcesI_(407D), and I_(408D)), and BLOCK_(402D) which is the minimum currentselector or MCS_(400D)(containing NMOSFET M_(421D), and PMOSFETsM_(420D), M_(422D), and M_(424D)). As stated earlier, AMP_(400D) iscomprised of amplifier gain stage of FIG. 4A coupled with BUF_(400D),where node 414A is connected to node 414D, node 415A is connected tonode 415D. AMP_(400D) is configured in unity gain with node 412A (theamplifier gain stage negative input terminal) connected to node 420D (orV_(400D))

Note that it would be obvious for one skilled in the art to utilize aICMA that is complementary version of ICMA_(400D) (i.e., PMOSFET basedor NMOSFET based). It would also be obvious for one skilled in the artto utilize an ICMA that is a complementary version of MCS_(400D) (i.e.,PMOSFET base or NMOSFETs based). Additionally, it would be obvious forone skilled in the art that plurality of ICMAs coupled with plurality ofMCSs can be utilized, which are complementary to one another (e.g.,NMOSFET based ICMA coupled with primarily PMOSFET based MCS, utilizedalong with PMOSFET based ICMA coupled with primarily NMOSFET based MCS).Moreover, it would be obvious for one skilled in the art to design abuffer driver that utilizes other equivalent functional implementationsthat are minimum current selector (MCS) or non-inverting current mirroramplifier (ICMA). For example, a loser take all (LTA) can perform theequivalent function of MCS. Another example would be to utilize, acurrent mirror amplifier (CMA) instead of an inverting current mirroramplifier (ICMA) by arranging the circuit signal signs. Another examplewould be to utilize a current amplifier to perform the equivalentfunction of a ICMA.

Describing the details of the circuit in FIG. 4D is as follows, settingaside the non-idealities (e.g., mismatches, offsets). Let's firstbriefly describe the operation of minimum current selector (MCS_(400D)),in the boundary conditions (with inputs near the rails), independent ofthe ICMA_(400D) for now. The drain currents ID_(M421D) and ID_(M424D)track the currents in FET_(400D), which are M_(431D) (sink NMOSFET) andM_(426D) (source PMOSFET), respectively. The output of MCS_(400D) isID_(424D). The function of MCS_(400D) is to select the min(ID_(M421D),ID_(M424D)). When vin1_(400D) and vin2_(400D) are both up near V_(DD),then M_(421D) (tracking M_(431D)) is hard on and M_(424D) (trackingM_(426D)) is off or nearly off. Here, ID_(M421D)>>i and ID_(M424D)<<i.Excluding role of ICMA_(400D) for now, if vin1_(400D)≈vin2_(400D) bothvoltages are near V_(DD)→MCS_(400D) operates and selects themin(ID_(M421D), ID_(M424D))≈ID_(M424D)<<i. Conversely, whenvin1_(400D)≈vin2_(400D) both voltages are near V_(SS), then M_(421D)(tracking M_(431D)) is off or near off and gate terminal of M_(424D)(tracking M_(426D)) tends to turn it hard on, but M_(424D) is deprivedof current since M_(422D) is off and blocking any current flow. Here,ID_(M421D)≈ID_(M420D)≈ID_(M422D)<<i. Again, excluding role ofICMA_(400D) for now, if vin1_(400D)≈vin2_(400D) both voltages are nearV_(SS)→MCS_(400D) operates and selects the min(ID_(M421D),ID_(M424D))≈ID_(M421D)<<i.

Before describing the BUF_(400D) operating in over drive (sink-sourcemodes) and steady state conditions, there is described the currentequilibrium conditions at nodes 414 and 415 for steady state to hold. Atnode 414D in FIG. 4D (which is connected to node 414A in FIG. 4A), KCLrequires the sum of ID_(M418A), ID_(M419A), I_(408D), and ID_(M429D) toconverge to zero for steady state. At node 415D in FIG. 4D (which isconnected to node 415A in FIG. 4A), KCL requires the sum of ID_(M416A),and ID_(M417A) to converge to zero for steady state. For AMP₄₀₀ unitygain loop (containing the amplifier gain stage of FIG. 4A and BUF_(400D)of FIG. 4D) to hold in steady state,(ID_(M417A)≈ID_(M419A))≈(ID_(M416A)≈ID_(M418A)) and I_(408D)≈ID_(M429D).This condition will be called “the conditions of current equilibrium fornodes 414D and 415D” in this section, which will be referred to later.

In the sink mode, when node 414D and node 415D are pulled up towardsV_(DD), MN_(431D) (i.e., sinking current for the external load) andMN_(421D) are turned on hard that causeID_(M421D)≈ID_(M420D)≈ID_(M422D)>>i. Here, M_(426D) and M_(424D) turnoff (initially) and ID_(M424D)<<i, which would block ID_(M422D). Thus,the output current of MCS_(400D) selects the min(ID_(M421D),ID_(M424D))=ID_(M424D)<<i, which flows onto M_(423D), andinitially→ID_(M423D)=ID_(M425D)<<i→all of I_(407D)=2i flows intoM_(427D)→ID_(M427D)≈ID_(M429D)=2i. For “conditions of currentequilibrium for nodes 414D and 415D” to hold, node 414D gets regulatedby being pulled down in part by ID_(M429D), that is initially beforeequilibrium runs at ‘2i’, until I_(408D)=i≈ID_(M429D). In summary, inorder for this current equilibrium to persist, ICMA_(400D) containedwithin AMP_(400D) loop regulate node 414D by lowering node 414D voltagesufficiently enough, which is when I_(408D)=iID_(M429D)→ID_(M424D)≈ID_(M423D)≈ID_(M425D)≈ID_(M427D)≈i≈ID_(M429D)≈i≈(1/s)×ID_(M426D).As such, for BUF_(400D) in current ‘sink mode’ whenID_(M431D)>>ID_(M426D), the inactive source FET, M_(426D), is regulatedto run at a controlled current ID_(426D)≈s×i. Note that ‘i’ is aconstant current source set by the main bias network of the amplifier,and ‘s’ is FET aspect ratio that is tightly controllable.

Conversely, in the source mode, when node 414D and 415D of FIG. 4D arepulled down towards V_(SS), and M_(426D) is turned on hard, whereinitially M_(431D), M_(421D), M_(420D), and M_(422D) are turned off orare near off. Although, gate terminal voltage of M_(424D) is pushed downtowards V_(SS), but M_(422D) being initially off or near off, starvesM_(424D). As such, MCS_(400D) initially selects the min(ID_(M421D),ID_(M424D))=ID_(M421D)<<i, that flows through M_(424D) and ontoM_(423D), which is the input ICMA_(400D). As such, before equilibrium,ID_(M421D)≈ID_(M420D)≈ID_(M422D)≈ID_(M424D)≈ID_(M423D)≈ID_(M425D)<<i→allof I_(407D)=2i would initially flow throughM_(427D)→ID_(M427D)≈ID_(M429D)≈2i>I_(408D)→creates a current imbalanceat node 414B that is unsustainable→AMP_(400D) loop kicks in to regulateback towards current equilibrium. In order for “conditions of currentequilibrium for nodes 414D and 415D” to return and hold, and to simplifythe explanation for clarity of description, in FIG. 4A(ID_(M416A)≈ID_(M418A)) can get very slightly increased initially above(ID_(M417A)≈ID_(M419A)) in FIG. 4A. Here, by AMP_(400D) loop regulation,the voltages at both node 414D and node 415D are both initially pulledup until M_(421D) (and its mirrors M_(431D)) turn back on sufficientlyenough for ID_(M421D)≈i, which brings current equilibrium. Note in FIG.4A that despite the very slight increase in (ID_(M416A)≈ID_(M418A)) over(ID_(M417A)≈ID_(M419A)) at node 414D, ICMA_(400D) (coupled and fed viaMCS_(400D)) have enough gain to regulate a slightly increased ID_(M429A)(compared to the fixed I_(408D)≈i) to keep the voltage at node 414D lowenough so that M_(426D) can continue sourcing current to the externalload. The only draw-back here shows up in AMP₄₀₀ DC (steady state) gainreduction as the sink-source currents for an external load is increased(i.e., the smaller the resistive load, the smaller the AMP₄₀₀ gain,which is not uncommon in inverting buffer drivers). Here, the“conditions of current equilibrium for nodes 414D and 415D” are held,and steady state conditions maintained, while M_(426D) to continuessourcing current. Just to capture the current balance and steady stateconditions here, note that withID_(M426D)>>ID_(M431D)→ID_(M424D)>>ID_(M421D)→min(ID_(M421D),ID_(M424D))=ID_(M421D)≈i→operating current in the inactive sink-FET,that is mirrored with M_(421D), is controlled and regulated atID_(M431D)≈k×i, while steady state condition hold and the source FET,M_(426D), continues sourcing extra current to an external load.

Again, ignoring non-idealities, here the mechanism for the BUF_(400D)operating in steady state is described, when there is no output load. Atsteady state conditions, I_(408D)≈ID_(M429D) at node 414D in FIG. 4D, aswell as in FIG. 4A where ID_(M418A)≈ID_(M419A) at node 414A andID_(M416A)≈ID_(M417A) at node 415A, andID_(M418A)≈ID_(M416A)≈ID_(M419A)≈ID_(M417A). In steady state, thecurrent output of the MCS_(400D) selects the min (ID_(M421D),ID_(M424D))≈ID_(M421A)≈ID_(M424A)≈i→I_(408D)≈ID_(M429D)≈ID_(M427D)≈ID_(M425D)≈ID_(M423D)≈i≈ID_(M424D).For clarification, note that the MCS_(400D) is arranged as a translinearcircuit. Here, the VG_(M420D)≈VG_(M424D) (relative to V_(DD)). WhenM_(420D) that has a W/L=1× and is in parallel with the seriesassociation of M_(424D)-M_(422D), each withW/L=2×→ID_(M424D)≈ID_(M422D)≈i≈ID_(M420)≈ID_(M421). Note thatVGS_(M420D)≈VGS_(M426D) and VGS_(M421D)≈VGS_(M431D). As such, by settingFET_(400D) scale factors k=s, the (inverting) output FETs currents, inthe buffers driver, operate with a controlled quiescent current ofID_(M431D)≈ID_(M426D)≈s×i while in steady state.

In summary, a near rail-to-rail input-output (i.e., inverting) bufferdriver is proposed utilizing a novel current mode output stagecomprising of minimum current selectors (MCS) and non-inverting currentmirror amplifiers (ICMA) which are fast and operate at low V_(DD). Thepreceding gain amplifier stage high-impedance high-gain node, dominatesin setting the AC and transient response, with minimal speed degradationcaused by the fast buffer driver. Such is the case in part because ofthe (inverting) buffer driver wide bandwidth since it operates chieflyin current mode makes its fast inherently. The (inverting) buffer driveris based on the main-stream, standard, low cost, and rugged standarddigital CMOS manufacturing platform.

In conclusion, the benefits of the FIG. 4D's BUF_(400D) comprisingMCS_(400D) and ICMA_(400D) include the following. First, the (inverting)buffer driver can work with minimum V_(DD) of V_(GS)+2V_(DS) (lowvoltage). Second, it principally operates in current mode which makes itinherently fast. Third, it is simple and made of a few transistor whichmakes it small, low cost, and higher quality. Fourth, it can run atultra low currents, which makes its power consumption ultra-low whencombined with its ability to perform to specifications with low powersupply voltage. Fifth, it can sink and source significantly highercurrents (compared with its own steady-state operating current) for anexternal load, and have regulated and controlled quiescent current inits inactive output FET which helps with the (inverting) buffer driversperformance, including more consistent DC and dynamic response, amongstother. FIG. 4G is a simulation of a buffer amplifier containing bufferdriver of FIG. 4D coupled with gain stage of FIG. 4A. This simulationsindicates approximate and typical sink-source current capability for thebuffer amplifier to drive external load resistors 900KΩ, 300KΩ, and30KΩ. The simulation indicates current consumption for the bufferamplifier of roughly 120 nA for FIG. 4D coupled with gain stage of FIG.4A. Sixth, the MCS that monitors the output FET's (BLOCK-400D) operatesin current mode. Also, the ICMA, while running chiefly in current mode,regulates the operating currents in the output (inactive) FETs. RunningMCS and ICMA chiefly in current mode enables them to be inherently fast,but additionally, processing signals in current mode reduces internalvoltage swings that helps with operating at lower voltages, which widensthe input-output voltage span closer to the rails. This is beneficial tolow power supply environments where there little signal to noise cushionavailable to waste for overhead at input or output terminals. Seventh,while utilization of current mirror inverter lowers the input impedanceat the internal inputs of ICMA that help with faster signal processingspeeds within ICMA. Moreover, the ICMA has a high output impedance that,when connected to an amplifier's gain stage, it neither substantiallyloads the amplifier's high impedance node nor does it dynamically slowthe amplifier. Eighth, (inverting) buffer driver needs no specialprocess and can be manufactured on rugged and time tested main-streamCMOS process that is inexpensive, readily available at many foundries,and with high quality. Ninth, this teaching can be tailored for(inverting) buffer drivers that need very high-speed, very high currentdrive capability, and near rail-to-rail input-outputs with low supplyvoltages s as well (i.e., can be very high current and very high speed).

The definitions of the words or elements of the claims shall include notonly the combination of elements which are literally set forth, but allequivalent structure, material or acts for performing substantially thesame function in substantially the same way to obtain substantially thesame result.

All references, including publications, patent applications, patents andwebsite content cited herein are hereby incorporated by reference to thesame extent as if each reference were individually and specificallyindicated to be incorporated by reference and was set forth in itsentirety herein.

The words used in this specification to describe the invention and itsvarious embodiments are to be understood not only in the sense of theircommonly defined meanings, but to include by special definition in thisspecification any structure, material or acts beyond the scope of thecommonly defined meanings. Thus if an element can be understood in thecontext of this specification as including more than one meaning, thenits use in a claim must be understood as being generic to all possiblemeanings supported by the specification and by the word itself.

Recitation of ranges of values herein are merely intended to serve as ashorthand method of referring individually to each separate valuefalling within the range, unless otherwise indicated herein, and eachseparate value is incorporated into the specification as if it wereindividually recited herein. Therefore, any given numerical range shallinclude whole and fractions of numbers within the range. For example,the range “1 to 10” shall be interpreted to specifically include wholenumbers between 1 and 10 (e.g., 1, 2, 3 . . . 9) and non-whole numbers(e.g., 1.1, 1.2, . . . 1.9).

Neither the Title (set forth at the beginning of the first page of thepresent application) nor the Abstract (set forth at the end of thepresent application) is to be taken as limiting in any way as the scopeof the disclosed invention(s). The title of the present application andheadings of sections provided in the present application are forconvenience only, and are not to be taken as limiting the disclosure inany way.

The invention claimed is:
 1. A method of regulating minimum operatingcurrent in a sink transistor and a source transistor of a buffer driverin an integrated circuit comprising: monitoring a sink transistorcurrent and a source transistor current in a buffer driver, wherein thebuffer driver is comprised of a sink transistor and a source transistor;detecting an imbalance between the sink transistor current and thesource transistor current by utilizing at least one of a minimum currentselect (MCS), a loser take all (LTA), a maximum current select (MXCS),and a winner take call (WTA) circuit; generating a regulation signal inresponse to detecting the imbalance between the sink transistor currentand the source transistor current by utilizing a current feedbackamplifier (CFA), wherein the CFA is comprised of at least one of acurrent mirror amplifier (CMA), an inverting CMA (ICMA), and anon-inverting CMA (NICMA); and utilizing the regulation signal toregulate a minimum operating current in at least one of the sinktransistor and the source transistor.
 2. The method of regulatingminimum operating current in a sink transistor and a source transistorof a buffer driver in an integrated circuit of claim 1, furthercomprising: rectifying the monitoring of the at least one of the sinktransistor current and the source transistor current.
 3. A buffer drivercircuit comprising: a buffer driver having a buffer driver input portand a buffer driver output port a driver field effect transistor (FET)having a driver FET input port and a driver FET output port, a currentimbalance detection (CID) circuit comprised of at least one of a minimumcurrent select (MCS), a loser take all (LTA), a maximum current select(MXCS), and a winner take all (WTA), wherein the CID circuit having aCID circuit input port and a CID circuit output port; a currentamplifier (CA) circuit comprised of at least one of a current mirroramplifier (CMA), an inverting CMA (ICMA), and a non-inverting CMA(NICMA), wherein the CA circuit having a CA circuit input port and a CAcircuit output port; wherein the driver FET output port communicateswith the buffer driver output port; wherein the driver FET input portcommunicates with the buffer driver input port; wherein the CID circuitinput port communicates with the driver FET input port, and the bufferdriver input port; wherein the CID circuit output port communicates withthe CA circuit input port; and wherein the CA circuit output portcommunicate with the driver FET, the CID circuit input port, and thebuffer driver input port.
 4. The buffer driver circuit of claim 3further comprising: a current rectifying (CR) circuit having a CRcircuit input and a CR circuit output port; wherein the CR circuit inputport communicates with the buffer driver input port, the CA circuit outport, and the input port of the driver FET; and wherein the CR circuitoutput port communicates with the input port of the CID circuit.